Oscillators – Automatic frequency stabilization using a phase or frequency... – Plural a.f.s. for a single oscillator
Reexamination Certificate
2002-09-06
2004-03-02
Mis, David C. (Department: 2817)
Oscillators
Automatic frequency stabilization using a phase or frequency...
Plural a.f.s. for a single oscillator
C331S00100A, C331S016000, C331S017000, C331S025000, C331S17700V, C331S03600C, C332S127000, C455S260000
Reexamination Certificate
active
06700447
ABSTRACT:
FIELD OF THE INVENTION
The invention relates to all areas where two-point phase modulation is used. In particular it relates to a Sigma Delta controlled fractional-N PLL modulator with an extra modulation input.
BACKGROUND OF THE INVENTION
A Sigma Delta controlled fractional-N PLL (phase-locked loop) is often used in a radio system for generating spurious-free local oscillator (LO)-frequencies and to allow faster frequency jumps. A block diagram of this is shown in
FIG. 1
By controlling a frequency divider
8
ratio with a sigma-delta modulator
9
, modulation with a constant envelope can be generated. By using these properties of the fractional N PLL
15
, compact radio architectures for constant envelope systems e.g., Global Sysem for Mobile Communications (GSM) or DCS could be developed. This also means that a complete radio could be integrated into a single ASIC (application specific integrated circuit).
Since constant envelope systems aren't bandwidth efficient, some proposed systems also use amplitude modulation. Examples of these systems are EDGE (enhanced data GSM environment) and WCDMA (wideband code division multiple access).
If the modulating signal is divided into a phase part and an amplitude part, the phase part could be introduced in the fractional N PLL and the amplitude part could be added (effectively multiplied) in an amplifier after the PLL. In this way, switching blocks could be used throughout the complete modulator, which is very power efficient.
But when dividing the signal into amplitude and phase part, the bandwidth of the phase and amplitude part become much wider than the combined signal. Since the amplitude and phase part are combined in a multiplier after the PLL
15
this imposes stringent requirements on dynamic range and bandwidth of the amplitude and phase part and also on the timing between the amplitude and phase part.
For the phase part this means that the usable modulation bandwidth of the fractional N PLL is not wide enough. This bandwidth is limited by PLL noise, sigma-delta noise and matching issues.
One way to get around this limitation is to add a second modulation point, i.e., two-point modulation. By also inserting the modulation signal after the PLL loop filter, there is no trade-off between PLL loop bandwidth and modulation bandwidth.
Referring now to the drawings, in
FIG. 1
, the instantaneous frequency first is inserted at two points
10
,
12
in the modulator and the amplitude part A(S) is inserted at the PA (power amplifier)
14
.
The transfer function from the modulation inputs to the output of the VCO (voltage controlled oscillator)
16
can be derived as
θ
out
,
VCO
(
s
)
=
f
inst
(
s
)
Ns
K
phd
K
vco
s
H
LP
(
s
)
1
+
K
phd
K
vco
s
H
LP
(
s
)
/
N
+
f
inst
(
s
)
K
vco
′
K
vco
s
1
+
K
phd
K
vco
s
H
LP
(
s
)
/
N
=
f
inst
(
s
)
s
K
vco
K
vco
′
+
K
phd
K
vco
s
H
LP
(
s
)
/
N
1
+
K
phd
K
vco
s
H
LP
(
s
)
/
N
=
[
If
,
K
vco
=
K
vco
′
]
=
f
inst
(
s
)
s
This is independent of PLL loop bandwidth.
This modulation scheme is included in patent document WO 99/07066, “Frequency synthesizer systems and methods for three point modulation with a DC-response.”
However, a new unknown is introduced, namely, the estimation of the VCO gain. If the estimation of the VCO gain is wrong we get spectral growth and break the ACPR (adjacent channel power ratio) requirement of the system.
In a RF-ASIC with onboard VCO the VCO-gain is dependent on the size of the inductor, the output frequency and the bias point of the varactor. A standard VCO configuration is depicted in FIG.
2
.
In
FIG. 2
, there are inductors L
1
, L
2
(
20
,
22
) and a parasitic capacitance (
24
) Cpar, a combination of all capacitor loading and all parasitics as seen from the resonator. Then there is the tuning network, consisting of Cc
26
,
28
(coupling capacitors to couple the varactors loosely to the resonator), Cv
30
,
32
(the actual varactors) and Rgnd
34
,
36
(ground reference for the varactors). The bottom part shows the active components (e.g., transistors
35
,
40
) responsive for sustaining the oscillation.
The tuning sensitivity of the VCO is derived by taking the derivative of the output frequency with respect to the tuning voltage.
w
o
=
1
L
tot
C
tot
;
∂
w
o
∂
V
tune
=
∂
w
o
∂
C
tot
∂
C
tot
∂
C
v
∂
C
v
∂
V
tune
=
-
L
tot
2
(
L
tot
C
tot
)
3
/
2
1
2
(
C
c
C
c
+
C
V
)
2
∂
C
V
∂
V
tune
=
-
L
tot
w
o
3
2
1
2
(
C
c
C
c
+
C
V
)
2
∂
C
V
∂
V
tune
The tuning sensitivity is dependent on many parameters.
The VCO on-chip inductor (e.g., L
1
, L
2
) is a large metal structure and is inherently stable.
The varactor capacitance and the slope of the varactor capacitance are dependent on the tuning voltage. The tuning voltage is dependent on the VCO center frequency.
By making a careful design keeping the above equation in mind, the total VCO gain variation could be reduced, but usually other parameters limit the flexibility.
A table with measured VCO gain versus frequency could compensate the variation. However, the main problem with this solution is that when manufacturing these circuits, the parasitic capacitance (Cpar) of the resonator varies and a different tuning voltage is required to get the correct output frequency. The VCO gain could vary as much as 50% from one sample to the other. This means that the VCO gain would have to be measured for each VCO chip to get stable performance.
In WO 99/07066 the author describes a different circuit configuration, where the VCO has two separate inputs, one for the PLL tuning voltage and one for the modulation input. A circuit configuration for this is depicted in FIG.
3
.
In
FIG. 3
, a separate tuning input V mod
50
is added for modulation. This input is similar to the original tune input V tune
42
but it has a DC-voltage applied to set the operating point of the varactor.
Using this modified VCO (
FIG. 3
) means that the varactors can be biased at a suitable DC-level. Also the input bandwidth and tuning sensitivity can be optimized for modulation. If the DC-level applied to the varactors is constant, the only thing varying in the previous formula is the center frequency. This solution is independent of parasitic capacitor variations, since this is compensated in the tuning voltage.
This means that the VCO-gain variation from sample to sample is mainly dependent on spread in the varactor at the specific bias point and spread in the coupling capacitor. By careful design this could be less than 10% (mainly by choosing large size components).
However, for future systems with more complex modulation schemes (for example 16 QAM) the requirement of VCO-gain estimations will be higher using this modulation scheme. Therefore, some kind of automatic calibration is necessary. Also, using automatic calibration would secure a higher yield and more stable operation.
SUMMARY OF THE INVENTION
The present invention overcomes the above limitations by means of a frequency synthesizer generating an output frequency. The frequency synthesizer comprises a controlled oscillator having a tuning input which is responsive to a frequency control input signal to generate an output frequency, and having a feed back loop. The frequency synthesizer further comprises a compensation circuit for compensating gain variation of the controlled oscillator outside of the feedback loop.
The feed back loop further comprises a phase comparator wherein the phase comparator is responsive to a reference frequency signal and a feedback signal for producing an error output signal. The phase comparator comprises a first charge pump and a second charge pump. The first charge pump receives the reference frequency signal and the second charge pump receives a divided signal. The first and second charge pumps produce first and second output currents, wherein
Jenkens & Gilchrist P.C.
Mis David C.
Telefonaktiebolaget LM Ericsson (Publ)
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