Wideband impedance coupler

Details Wideband impedance coupler Wideband impedance coupler

2001-09-20

2003-10-28

Tokar, Michael (Department: 2819)

Wave transmission lines and networks

Coupling networks

With impedance matching

C333S238000

Reexamination Certificate

active

06639487

ABSTRACT:

PRIORITY CLAIM
This is a national stage of PCT application No. PCT/FI00/00066, filed on Feb. 1, 2000. Priority is claimed on that application.
FIELD OF THE INVENTION
The invention relates to a method for matching the characteristic impedances of a transmission line when the transmission line is taken into a wall made of dielectric material. The invention also relates to a transmission line characteristic impedance coupler to change the characteristic impedance of a transmission line.
BACKGROUND OF THE INVENTION
In certain RF structures, a signal transmission line has to be modified in terms of either dimensions or structure. One such case is a signal line feedthrough from free space into a hermetically sealed Monolithic Microwave Integrated Circuit (NIMIC) integrated circuit package. When such a feedthrough is realized in the wall of the package, the characteristic impedance changes at the interfaces of the feedthrough. That change i: caused by the change in the conductor structure, the change in the relative permittivity (&egr;
r
) of the material around the conductor at tie interface, and by possible ground potential planes in the vicinity of the conductor. These factors together affect the shape of the electromagnetic field on tie different sides of the interface. The change in the field shape causes part of the signal arriving at the interface to be reflected back in its direction of incidence. The ratio of the reflected signal to the signal incident upon the interface, designated as either &rgr; or, commonly in RF technology, as S
11
, return attenuation, is obtained from equation (1). The smaller ratio, the better the matching of the characteristic impedance at the interface of the feedthrough.
S
11
=
Z
2
-
Z
1
Z
2
+
Z
1
,
(
1
)
where
S
11
=reflection coefficient,
Z
1
=characteristic impedance of the conductor coming to the interface,
Z
2
=characteristic impedance of the conductor leaving the interface.
This power loss at the interface caused by a mismatch of characteristic impedances is called reflection attenuation, equation (2).
Γ
=
10
⁢
 
⁢
lg
⁢
1
1
-
&LeftBracketingBar;
S
11
&RightBracketingBar;
2
⁡
[
dB
]
,
(
2
)
where
&Ggr;is the reflection attenuation in decibels.
In practice, the magnitude of the return attenuation is strongly dependent of the frequency used and, thereby, its degradation limits the frequency range desired by the user.
Another problem caused by an interface is the insertion loss occurring at the interface. In RF technology, it is often referred to by parameter S
21
. Its magnitude depends on the radiation losses at the interface, reflection attenuation and the different relative permittivities (&egr;
r
) of the materials on the different sides of the interface. Insertion loss also depends strongly on the frequency used since the permittivities (&egr;
r
) of materials change as the frequency becomes higher. Minimization of insertion losses is just as important as the minimization of the return attenuation in the desired frequency band if one wants to achieve good and low-loss transmission path matching at the interface.
Signal transmission paths in RF applications generally consist of coaxial conductors, striplines, microstrip conductors or coplanar conductors in various combinations. When looking for conductors that do not require much space or that can be planted on a substrate, one chooses either microstrip or coplanar conductors. The advantage of these conductors compared e.g. to coaxial cable is that they can be realized planar as far as the signal conductors are concerned. In the coplanar conductor structure, also the so-called ground conductor may be realized in the same plane with the signal conductor proper.
One way of matching the transmission line at the interface is to use a quarter-wave transformer shown in
FIG. 1
a
, based on changing the width of the conductor in steps of &lgr;/4. A conductor
101
is placed on a suitable substrate
102
. The width of the conductor is changed in four steps
103
. However, the matching achieved in this way works only for a relatively narrow frequency band. The cause of this is the discontinuity that occurs at the steps
103
, causing unwanted reactive fields or radiation into space at said steps
103
.
Another widely used matching technique is so-called tapering. It means that the geometry of a conductor is changed by tapering it continuously for ½ to 1 &lgr; from original dimensions to desired dimensions, as shown in
FIG. 1
b
. A conductor
104
is placed on a substrate
102
. Tapering
105
of the conductor is realized without steps, i.e. continuously. Characteristic impedance matching realized by means of tapering is more controlled than impedance matching based on a quarter-wave transformer. Thus the unwanted phenomena occurring at the interface are smaller and the various losses will not increase together with the frequency as strongly as with a quarter-wave transformer.
In the publication “IEEE Transactions on Components, Packaging and Manufacturing Technology—Part B, vol 20, No. 1 February 1997, Decker & al, Multichip MMIC Package for X and Ka Band” there is presented a solution for realizing a more wideband matching for a feedthrough in a MMIC package. In that solution the transmission line matching is realized by tapering the conductor before taking it inside the MMIC package. The material of the wall of the MMIC package is an insulator the relative permittivity (&egr;
r
) of which is greater than the relative permittivity (&egr;
r
) of air.
FIG. 2
illustrates the principle of the coupler arrangement thus realized. On top of the base structure
203
of the package there is a continuous ground plane
202
made of conductive material. On top of the ground plane there is a substrate
201
made of insulating material, and on top of the substrate there is a coplanar conductor structure, a signal conductor
204
and ground conductors
205
. Near the conductors in the feedthrough there are also ground planes
206
which are connected through vias
209
to the ground plane under the substrate. The wall
208
of the package is made of insulating material as well. The characteristic impedance of the coplanar conductor changes as the conductors are taken into the wall of the package. Matching for the impedance change is realized by tapering
207
. As seen from
FIG. 2
, tapering of the conductor is realized before the conductor is taken into the insulating material that the package walls consist of. Likewise, when the conductors come out of the wall material, another tapering
210
is realized which, too, is realized in free space. The feedthrough in the wall of a MMIC package according to this solution is applicable at up to 26 GHz, but not in the Ka band.
The return attenuation of the MMIC package feedthrough solution presented in the referenced document stays below −15 dB at up to 27.5 GHz. The insertion attenuation is of the order of 1 dB at up to 30 GHz, whereafter it grows rapidly.
In the publication Ishitsuka, T and Sato, N, Low Cost High-Performance Package for a Multi-Chip MMIC Modules, GaAs Symp. Dig. November 1988, pp. 221-224, there is presented another solution for a signal conductor feedthrough in a MMIC package. In that solution, the walls
208
of the MMIC package are comprised of multilayer ceramic sheets metallized on both sides. The ground potential planes resulting in the different layers are interconnected through several vias
209
. The structure of the feedthrough of the signal conductor proper is otherwise like that described in the previously referenced document. This structure stretches the useable frequency band up to the 30 GHz limit. Disadvantages include the complexity of the wall structure and the resulting expensiveness of the structure.
The structures described in the publications mentioned above often employ GaAs-based chips. In GaAs ICs the coupling points of the signal conductors are located on the upper surface of the microchip, and the lower surface is covered by a continuous ground plane. When conduct

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