Communications: radio wave antennas – Antennas – With radio cabinet
Reexamination Certificate
1999-11-09
2002-07-02
Wong, Don (Department: 2821)
Communications: radio wave antennas
Antennas
With radio cabinet
C333S126000, C333S208000
Reexamination Certificate
active
06414639
ABSTRACT:
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a resonance device in which a transmission line such as a micro-strip line or a coplanar line is coupled to a resonator. In addition, the invention relates to an oscillator, a filter, a duplexer, and a communication device incorporating the same.
2. Description of the Related Art
A conventional resonance device will be illustrated referring to FIG.
12
. This figure is a perspective view of the conventional resonance device.
The conventional resonance device
110
shown in
FIG. 12
is constituted of a micro-strip line
120
as a transmission line and a resonator
111
. The micro-strip line
120
is composed of a dielectric substrate
121
, a main conductor
122
formed on the upper surface thereof and an earth conductor
123
formed on the lower surface thereof. The resonator
111
is a cylindrical dielectric member, a part of which is arranged over the main conductor
122
of the micro-strip line
120
. In the resonance device
110
having such a structure, an electromagnetic field is excited surrounding the micro-strip line
120
by current flowing through the main conductor
122
of the micro-strip line
120
. As a result, the electromagnetic field excited by the current is coupled to the resonator
111
so that the resonator
111
resonates in a TE
01
&dgr; mode.
In general, when a resonance device is used to form an oscillator or a filter, a part of the characteristics of the oscillator or the filter depends on the strength of the coupling between a transmission line and a resonator used in the resonance device. For example, the stronger the coupling between the transmission line and the resonator, the greater the oscillating output of the oscillator, and the wider the band width characteristics of the filter.
In such a conventional resonance device, however, coupling beyond a certain level of strength cannot be obtained due to the dispersive characteristics of a micro-strip line, which will be described below. The dispersive characteristics of a micro-strip line are also described in “Microwave Planar Passive Circuits and Filters,” by J. Helszajn, John Wiley & Sons, 1994, pp 90-93, and other publications. Thus, when an oscillator having a large output and a-filter having wide frequency bandwidth characteristics are desired, since it is impossible to make the coupling between the transmission line and the resonator stronger than a certain level, there is a problem in that an oscillator and a filter having such desired characteristics cannot be obtained.
Referring to
FIG. 13
, a description will be given of the problem.
FIG. 13
is a graph showing the result of a simulation about the reflection characteristics of a resonance device with respect to a frequency. In this figure, reference numerals S
11
indicates the value of reflection characteristics, which is a ratio of output-signal strength/input-signal strength obtained when a signal is input from one side of a micro-strip line shown in FIG.
12
and an output signal is observed on the same side.
The resonance device used in the simulation has a structure shown in
FIG. 12
, in which the relative permittivity of the dielectric substrate
121
of the micro-strip line
120
is set to be 3.2, the thickness of the dielectric substrate
121
is set to be 0.3 mm, and the line width of the main conductor
122
is set to be 0.72 mm. In addition, the relative permittivity of the resonator
111
is set to be
24
, the diameter thereof is set to be 2.0 mm, and the thickness thereof is set to be 0.8 mm. As indicated by the graph shown in
FIG. 13
, in the conventional resonance device
110
, the reflection characteristics is 3 dB when the resonating frequency is 28.5 GHz. In other words, this shows a fact that in the case of such a conventional resonance device, many signals pass through without being reflected at a resonance frequency, with an implication that coupling between the micro-strip line
12
and the resonator
111
in the resonance device
110
is weak.
A description will be given below about the reason why the coupling between the transmission line and the resonator in the conventional resonance device is weak. This is a case in which a micro-strip line is used as the transmission line.
In general, in a micro-strip line, it is ideal that an electromagnetic field excited by current flowing through a main conductor all exists on a surface vertical to a signal-propagating direction. However, in fact, an electromagnetic field is distributed both in an air space around the micro-strip line and in a dielectric substrate. Since the permittivity of the air space and that of the dielectric substrate are different, a phase velocity of the electromagnetic field is different between the air space and the dielectric substrate. As a result, it is impossible to obtain the ideal situation in which the electromagnetic field all exists on the surface vertical to a signal-propagating direction. That is, in this situation, the electromagnetic field excited by current flowing through the main conductor includes a component parallel to a signal-propagating direction.
FIGS. 14A and 14B
each show the distribution of the electromagnetic field having the component parallel to a signal-propagating direction.
FIG. 14A
shows the distribution of an electric field and
FIG. 14B
shows that of a magnetic field.
According to an equivalent principle, in the conventional resonance device, the electromagnetic field associated with coupling between the resonator and the transmission line is an electromagnetic field in a direction substantially vertical to a signal-propagating direction. In contrast, the electromagnetic field in a direction parallel thereto is not associated with coupling between the resonator and the transmission line. In other words, when the electromagnetic-field component parallel to a signal-propagating direction is increased, it is suggested that this increases an undesired electromagnetic-field component in terms of the coupling between the resonator and the transmission line. Thus, this is a factor that weakens the coupling between them.
Meanwhile, the higher the frequency, the larger the electromagnetic-field component parallel to a signal-propagating direction. This will be described referring to
FIG. 15
, which shows the relationship between an effective relative permittivity and a frequency. In addition, a micro-strip line used in this situation has a structure shown in
FIG. 12
, in which the relative permittivity of the dielectric substrate
121
is set to be 3.2, the thickness of the dielectric substrate
121
is set to be 0.3 mm, and the line width of the main conductor
122
is set to be 0.72 mm.
In the micro-strip line shown in
FIG. 12
, as described above, although the electromagnetic field is distributed both in the air space around the micro-strip line and in the dielectric substrate, the permittivity of the air space is different from that of the dielectric substrate. As a result, energy existing in the air space flows into the dielectric substrate by which distortion occurs in the distributions of the electromagnetic field, with the result that an electromagnetic-field component parallel to a signal-propagating direction is generated. In other words, the higher the proportional amount of energy existing in the dielectric substrate, the larger the electromagnetic field-component parallel to a signal-propagating direction, which weakens coupling between the resonator and the transmission line.
Next, a description will be given of the relationship between the ratio of the amount of energy existing in the dielectric substrate and an effective relative permittivity. For example, when the relative permittivity of the dielectric substrate is indicated by the symbol er and the ratio between the energy existing in the air space and that in the dielectric substrate is set to 1:1, the effective relative permittivity indicated by the symbol e eff is approximately equal to (1+&egr;r)/2. When the energy existing in the dielectric
Chen Shih-Chao
Murata Manufacturing Co. Ltd.
Ostrolenk Faber Gerb & Soffen, LLP
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