Pulse or digital communications – Receivers – Automatic frequency control
Reexamination Certificate
2000-11-15
2004-09-28
Chin, Stephen (Department: 2634)
Pulse or digital communications
Receivers
Automatic frequency control
C455S164100, C455S192200
Reexamination Certificate
active
06798853
ABSTRACT:
BACKGROUND
The present invention relates to systems for providing automatic frequency control in radio frequency communication systems and, more specifically, to a system for compensating for frequency offset in a digital mobile radio frequency communication system through automatic frequency control.
The cellular telephone industry has made phenomenal strides in commercial operations in the United States as well as the rest of the world. Growth in major metropolitan areas has far exceeded expectations and is rapidly outstripping system capacity. If this trend continues, the effects of this industry's growth will soon reach even the smallest markets. Innovative solutions are required to meet these increasing capacity needs as well as maintain high quality service and avoid rising prices.
Digital communication systems, at their most basic level, provide for the transmission and reception of electronic messages between and among communication partners. The transmissions are effected through transmitters that modulate or encode the message and transmit the message in analog form for passage across a channel. At the receiver, the analog signal is converted back to the digital data of the message. Although paired transmitters and receivers are assigned to the same carrier, and the receiver is designed to perfectly demodulate (or decode) the modulated, transmitted signal, frequency offsets, or deviations, in the received signal may occur because of imperfections of oscillators and frequency synthesizers in the receiver. The frequency offset becomes, with time, a growing phase drift, which compromises the ability of the receiver to accurately and efficiently receive the transmitted messages. Therefore, in order to accurately detect sent information with minimal reception performance loss, the frequency offset of the received signal should be taken into consideration in the receiver design and compensated for during equalization.
Within the standards set for mobile radio frequency (“RF”) communication systems, frequency offsets of up to several hundred hertz are allowed. For example, a system conforming to the Global System for Mobile Communication (“GSM”) has a channel spacing of 200 KHz, thereby providing some tolerance for frequency offset without encountering the risk of receiving the wrong channel of transmitted data or receiving the transmitted data incorrectly. In contrast, however, a digital satellite communications system may use a channel spacing of only 5 KHz, with a correspondingly tight tolerance for frequency offset of received data.
In digital cellular telephones, automatic frequency control (“AFC”) is commonly used in RF receivers to keep the receiver locked on a particular frequency, despite imperfect component stability that would otherwise result in frequency drift. In contemporary digital communication systems, AFC is commonly based on second order digital phase-locked loop (“PLL”) filters that implement phase offset compensators to enable reliable communications. Such conventional AFC systems are used, for example, in current ANSI-136 systems, and effectively attempt to determine the phase error, to eliminate the phase drift. For a more thorough discussion of PLL filters and their use to determine phase error, see W. Lindsey and C. Chie, “A Survey of Digital Phase-Locked Loops,” 69 Proc. IEEE 410-31 (April 1981); K. J. Molnar and G. E. Bottomley, “Adaptive Array Processing MLSE Receivers for TDMA Digital Cellular/PCS Communications,” 16 IEEE J. Selected Areas in Comm. 1340-51 (October 1998).
More specifically, the impact of phase drift on an actual sampled, received data signal r(n) as a function of time index n=1, 2, 3, . . . , is commonly modeled as:
r
(
n
)
=e
j&phgr;(n)
z
(
n
)
+v
(
n
) (1)
where v(n) is additive noise from the transmission, and z(n) represents the desired signal, i.e., the signal carrying the transmitted data, and is represented by:
z
(
n
)
=
∑
k
=
0
L
-
1
h
(
k
)
s
(
n
-
k
)
,
(
2
)
where h(k) is a set of channel estimates for the channel across which the signals have been transmitted and L is the number of taps for the channel, and where s(n−k) is the transmitted signal associated with the time index (n−k). &phgr;(n) of equation (1) is the phase drift, given by:
&phgr;(
n
)=(
n−n
0
)&ohgr;
0
, (3)
with &ohgr;
0
being the frequency offset and n
0
being the reference sample data index, defining the sample position where &phgr;(n
0
)=0. More basically, the phase drift can be viewed as the frequency offset multiplied by time. This phase drift will be added, if uncorrected, to the phase of the desired signal, z(n). By estimating the phase drift as &phgr;′(n), the received signal can be phase compensated as:
ŕ
(
n
)=
r
(
n
)
e
−j&phgr;′(n)
(4)
If the estimate of the phase drift is close to &phgr;(n), and if the signal-to-noise ratio is sufficiently high, then the resultant signal ŕ(n) should be a good signal from which to detect the transmitted data. On the other hand, if nothing is done to compensate for errors in the estimates of the phase drift, the phase errors will degrade the ability of the receiver to determine the transmitted data. The detection of the sent data is thus based on phase-compensated data signals, with the aim of the AFC to provide accurate estimates of the phase drift. Conventional digital AFC systems provide this compensation as follows:
At each increment of the time index n, a phase error estimate is first calculated according to:
&phgr;
error
(
n
)=
arg
(
e
j&phgr;′(n)
r
*(
n
)
ź
(
n
)) (5)
where arg( ) denotes phase, * denotes conjugation, ź(n) denotes an estimate of the desired signal z(n), and &phgr;′(n) is an estimate of the phase drift &phgr;(n). Then, using the calculated phase error estimate, a new estimate of the frequency offset and a new estimate of the phase drift are determined by a second order filter according to:
{circumflex over (&ohgr;)}
0
(
n+
1)={circumflex over (&ohgr;)}
0
(
n
)+
K
1
&phgr;
error
(
n
) (6a)
&phgr;′(
n+
1)=&phgr;′(
n
)+{circumflex over (&ohgr;)}
0
(
n+
1)+
K
2
&phgr;
error
(
n
) (6b)
in which K
1
and K
2
are two constant filter parameters, where {circumflex over (&ohgr;)}
0
(n+1) denotes an updated estimate of the frequency offset &ohgr;
0
, and where. &phgr;′(n+1) denotes an updated estimate of the phase drift. This scheme requires the estimation of the desired signal, or ź(n). The choice of parameters K
1
and K
2
is a trade-off between fast convergence to the true frequency offset and sensitivity to noise. The parameters are set by prior simulations of data transmissions, using an upper estimate of the frequency offset, &ohgr;
0
, and an estimate of the anticipated noise, v(n). For example, when considering symbol spaced sampled received signals in a GSM system, typical values for K
1
and K
2
are less than 0.05 and 0.15, respectively.
Referring now to
FIG. 1
, there is shown a block diagram of an example of an automatic frequency control system. A data signal, r(n), received across a Channel
100
is directed to a Detector
102
to determine the transmitted data. The Detector
102
includes, for example, a Channel Estimator
104
, an Equalizer
106
, and an AFC
108
. The received data signal, r(n), is directed to the Channel Estimator
104
, where values of h(k) (channel estimates) of equation (2) are determined by comparing the training sequence within each received burst of data against the known data sequence that corresponds thereto (see also
FIG. 3
b
). The received data signal, r(n), is also directed to both the Equalizer
106
and the AFC
108
. The Equalizer
106
produces the desired signal estimate, ź(n), for input into the AFC
108
, and the soft output data, for input into the Decoder
110
. The desired signal estimate, ź(n), is input to the
Lindbom Lars
Zangi Kambiz
Chin Stephen
Kim Kevin
Telefonaktiebolaget LM Ericsson (publ)
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