Electric power conversion systems – Current conversion – Including d.c.-a.c.-d.c. converter
Reexamination Certificate
2001-05-15
2002-04-02
Berhane, Adolf Deneke (Department: 2838)
Electric power conversion systems
Current conversion
Including d.c.-a.c.-d.c. converter
C363S021040, C363S097000
Reexamination Certificate
active
06366476
ABSTRACT:
BACKGROUND OF THE INVENTION
The present invention relates to a switching power supply circuit to be provided as a power supply for various electronic apparatus.
Switching power supply circuits employing switching converters such as flyback converters and forward converters are widely known. These switching converters form a rectangular waveform in switching operation, and therefore there is a limit to suppression of switching noise. It is also known that because of their operating characteristics, there is a limit to improvement of power conversion efficiency.
Hence, there have been proposed various switching power supply circuits formed by various resonance type converters that make it possible to readily obtain high power conversion efficiency and to achieve low noise by forming a sinusoidal waveform in switching operation. The resonance type converters have another advantage of being able to be formed by a relatively small number of parts.
FIGS. 9 and 10
are circuit diagrams each showing an example of a prior art switching power supply circuit employing a resonance type converter.
This voltage resonance type converter is externally excited, and a MOS-FET, for example, is used as a switching device Q
1
.
A capacitor Cr is connected in parallel with a drain and a source of the switching device Q
1
. Capacitance of the capacitor Cr and leakage inductance obtained at a primary winding N
1
of an isolating converter transformer PIT form a voltage resonant circuit. The parallel resonant circuit performs resonant operation according to switching operation of the switching device Q
1
.
A clamp diode (so-called body diode) DD is connected in parallel with the drain and source of the switching device Q
1
. The clamp diode DD forms a path of clamp current that flows during an off period of the switching device.
The drain of the switching device Q
1
is connected to an oscillating circuit
11
in a switching driver
10
. An output of the drain supplied to the oscillating circuit
11
is used to variably control an on period of switching operation of the switching device Q
1
in switching frequency control.
The switching device Q
1
is driven for switching operation by the switching driver
10
which is formed by integrating the oscillating circuit
11
and a driving circuit
12
, and the switching frequency of the switching device Q
1
is variably controlled for the purpose of constant-voltage control. Incidentally, the switching driver
10
in this case is provided as a single integrated circuit, for example.
The switching driver
10
is connected to a line of rectified and smoothed voltage Ei via a starting resistance Rs. The switching driver
10
starts operation by being supplied with power supply voltage via the starting resistance Rs at the start of power supply, for example.
The oscillating circuit
11
in the switching driver
10
performs oscillating operation to generate and output an oscillating signal. The driving circuit
12
converts the oscillating signal into a driving voltage, and then outputs the driving voltage to a gate of the switching device Q
1
. Thus, the switching device Q
1
performs switching operation according to the oscillating signal generated by the oscillating circuit
11
. Therefore, the switching frequency and duty ratio of an on/off period within one switching cycle of the switching device Q
1
are determined depending on the oscillating signal generated by the oscillating circuit
11
.
The oscillating circuit
11
changes frequency fs of the oscillating signal on the basis of the level of a secondary-side direct-current output voltage E
0
inputted via a photocoupler
30
. The oscillating circuit
11
changes the switching frequency fs and at the same time, controls the waveform of the oscillating signal in such a manner that a period TOFF during which the switching device Q
1
is turned off is fixed and a period TON during which the switching device Q
1
is turned on is changed. The period TON is variably controlled on the basis of the level of a switching resonance pulse voltage V
1
across the parallel resonant capacitor Cr.
As a result of such operation of the oscillating circuit
11
, the secondary-side direct-current output voltage E
0
is stabilized.
The isolating converter transformer PIT transmits switching output of the switching device Q
1
to the secondary side of the switching power supply circuit.
As shown in
FIG. 11
, the isolating converter transformer PIT has an E-E-shaped core formed by combining E-shaped cores CR
1
and CR
2
made for example of a ferrite material in such a manner that magnetic legs of the core CR
1
are opposed to magnetic legs of the core CR
2
. A gap G is formed in a central magnetic leg of the E-E-shaped core in a manner as shown in the figure, and a primary winding N
1
and a secondary winding N
2
are wound around the central magnetic leg in a state in which the windings are divided from each other by using a dividing bobbin B. Thus, a state of loose coupling at a required coupling coefficient, for example k≈0.85 is obtained between the primary winding N
1
and the secondary winding N
2
, and because of the looseness of the coupling, a saturated state is not readily obtained.
The gap G can be formed by making the central magnetic leg of each of the E-shaped cores CR
1
and CR
2
shorter than two outer legs of each of the E-shaped cores CR
1
and CR
2
.
As shown in
FIGS. 9 and 10
, an ending point of the primary winding N
1
of the isolating converter transformer PIT is connected to the drain of the switching device Q
1
, while a starting point of the primary winding N
1
is connected to the rectified and smoothed voltage Ei. Hence, the primary winding N
1
is supplied with the switching output of the switching device Q
1
, whereby an alternating voltage whose cycle corresponds to the switching frequency of the switching device Q
1
occurs in the primary winding N
1
.
The alternating voltage induced by the primary winding N
1
occurs in the secondary winding N
2
on the secondary side of the isolating converter transformer PIT. In
FIG. 9
, a secondary-side parallel resonant capacitor C
2
is connected in parallel with the secondary winding N
2
, and in
FIG. 10
, a secondary-side series resonant capacitor C
2
is connected in series with the secondary winding N
2
. Therefore leakage inductance L
2
of the secondary winding N
2
and capacitance of the secondary-side parallel or series resonant capacitor C
2
form a resonant circuit. The resonant circuit converts the alternating voltage induced in the secondary winding N
2
into a resonance voltage, whereby voltage resonance operation is obtained on the secondary side.
The power supply circuit is provided with the parallel resonant circuit to convert switching operation into voltage resonance type operation on the primary side, and the parallel or series resonant circuit to provide voltage resonance operation on the secondary side. In the present specification, the switching converter provided with resonant circuits on the primary side and the secondary side as described above is referred to as a “complex resonance type switching converter.”
A rectifying and smoothing circuit comprising a bridge rectifier circuit DBR and a smoothing capacitor C
0
is provided on the secondary side of the power supply circuit, whereby a secondary-side direct-current output voltage E
0
is obtained. In the power supply circuit of
FIG. 9
, full-wave rectifying operation is provided by the bridge rectifier circuit DBR on the secondary side. In this case, the bridge rectifier circuit DBR is supplied with the resonance voltage by the secondary-side parallel resonant circuit, and then generates the secondary-side direct-current output voltage E
0
whose level is substantially equal to that of the alternating voltage induced in the secondary winding N
2
. In the power supply circuit of
FIG. 10
, two rectifier diodes D
01
and D
02
are connected in a manner shown in the figure, and therefore the rectifier circuit on the secondary side forms a voltage doubler rectifier circuit
Berhane Adolf Deneke
Maioli Jay H.
Sony Corporation
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