Communications: directive radio wave systems and devices (e.g. – Return signal controls external device – Missile or spacecraft guidance
Reexamination Certificate
2002-04-02
2004-06-22
Tarcza, Thomas H. (Department: 3662)
Communications: directive radio wave systems and devices (e.g.,
Return signal controls external device
Missile or spacecraft guidance
C342S021000, C342S098000, C342S109000, C342S192000
Reexamination Certificate
active
06753803
ABSTRACT:
FIELD OF THE INVENTION
This invention relates to a method and apparatus for detecting signals. The invention is especially, but not exclusively, applicable to systems which may be subject to an unknown Doppler frequency shift affecting modulated coherent signals used for communication and ranging purposes, and particularly substantially continuous signals modulated by random or chaotic waveforms utilized by obstacle detection or collision avoidance systems.
BACKGROUND OF THE INVENTION
Such systems are designed to operate in multiuser and often hostile environments, and are intended for a wide range of applications, including automotive applications, industrial robotics and unmanned vehicle navigation.
FIG. 1
is a block diagram of a typical microwave obstacle-detection system. The system comprises a signal generator
1
that produces a substantially continuous waveform x(t) with suitable bandwidth to provide required range resolution. The waveform x(t) may be deterministic (periodic or aperiodic), chaotic or purely random.
The system also has a microwave oscillator
2
that generates a sinusoidal signal with required carrier frequency, a modulator
3
that modulates one or more of the parameters (such as amplitude, phase, or frequency) of the carrier signal with the modulating waveform x(t), a power amplifier (PA)
4
that amplifies the modulated carrier signal to a required level, a microwave transmit antenna (TA)
5
that radiates an electromagnetic wave representing the modulated carrier signal towards an obstacle
6
, a microwave receive antenna (RA)
7
that receives an electromagnetic wave reflected back by the obstacle
6
, an input amplifier (IA)
8
that amplifies a signal provided by the receive antenna (RA)
7
, and a coherent demodulator
9
that processes jointly the reference carrier signal supplied by the oscillator
2
and a signal supplied by the input amplifier (IA)
8
to reconstruct a time-delayed replica y(t) of the modulating waveform x(t).
The modulating waveform x(t) and its time-delayed replica y(t) are then processed jointly during a specified time interval by a suitable processor
10
, such as correlator, to produce an estimate of the unknown time delay that is proportional to the distance (range) between the system and the obstacle
6
.
FIG. 2
shows an example of a correlation function of a synchronous random binary waveform.
When there occurs a relative movement between a ranging system and an obstacle of interest, an electromagnetic wave reflected back by the obstacle and received by a coherent system will exhibit a Doppler frequency shift. The value &ohgr;
D0
of this (angular) frequency shift can be determined from:
ω
D0
=
2
υ
0
c
ω
0
where &ngr;
0
is the radial speed (i.e., the range rate) of a relative movement between the system and the obstacle and &ohgr;
0
is the (angular) carrier frequency of a transmitted electromagnetic wave having velocity c.
The signal reflected back by a moving obstacle can be expressed as:
z
r
(
t
)=
&ggr;x
(
t−&tgr;
0
)cos [(&ohgr;
0
+&ohgr;
D0
)(
t−&tgr;
0
)+&thgr;]
where &ggr; is the round-trip attenuation, &tgr;
0
is the delay corresponding to the range D, &ohgr;
D0
is the Doppler frequency shift, and &thgr; is an unknown constant phase shift. Strictly speaking, the value of &tgr;
0
cannot be constant for a nonzero Doppler frequency &ohgr;
D0
. However, in most practical cases it is assumed that the small changes in range D, of the order of the wavelength of the carrier frequency, cannot be discerned when determining a short-time time-delay estimate. Such assumption justifies decoupling delay and Doppler frequency measurements.
A baseband signal corresponding to the received signal z
r
(t) has the form
y
(
t
)=
x
(
t−&tgr;
0
)cos(&ohgr;
D0
t+&phgr;
)
where &phgr; is an unknown phase shift.
The correlator performs the following operation
R
x
y
(
τ
)
=
1
T
0
∫
0
T
0
x
(
t
-
τ
)
x
(
t
-
τ
0
)
cos
(
ω
D0
t
+
ϕ
)
ⅆ
t
where the integral is evaluated for a plurality of hypothesized time delays &tgr;
min
<&tgr;<&tgr;
max
. When the observation interval T
0
is much shorter than one period (2&pgr;/&ohgr;
D0
) of the Doppler frequency, the value of cos(&ohgr;
D0
t
m
+&phgr;) is almost constant during T
0
, so that the correlation integral can be approximated by
R
x
y
(
τ
)
≈
1
T
0
cos
(
ω
DO
t
m
+
ϕ
)
∫
0
T
0
x
(
t
-
τ
)
x
(
t
-
τ
0
)
ⅆ
t
where the time instant t
m
is taken in the middle of the observation interval T
0
.
When the correlation integral is calculated repeatedly for successive short processing intervals, each of duration T
0
, the sequence of observed correlation functions may be represented by the plot of FIG.
3
. The rate of change (in time) of the correlation function R
xy
(&tgr;) will correspond to the Doppler frequency &ohgr;
D0
. The value of this frequency can be determined by applying some suitable form of spectral analysis.
FIG. 4
is a block diagram of a conventional correlator, comprising variable delay line
11
, multiplier
12
and integrator
13
, followed by a spectrum analyser
14
. When the total number of successive short processing intervals is large enough, the frequency spectrum S(&ohgr;), observed at the output of the analyser, will exhibit a pronounced peak at the Doppler frequency &ohgr;
D0
.
FIG. 5
is a block diagram of a multichannel correlator that uses a tapped delay line
15
to cover the entire interval of hypothesized time delays, &tgr;
min
<&tgr;<&tgr;
max
, in J steps of unit delay &Dgr; using delay circuits
16
. Multipliers
17
process the delayed signals wiith y(t) and provide the outputs to integrators
18
. The required spectral analysis may be performed by a digital processor
19
implementing the Discrete Fourier Transform (DFT). The principle of operation is thus similar to that of FIG.
4
.
The systems shown in
FIGS. 4 and 5
, and also other similar known systems, attempt to approximate the correlation integral with two parameters of interest, delay time &tgr; and Doppler frequency &ohgr;
D
by combining a time sequence of correlation functions determined over a number of relatively short observation intervals T
0
. Because of this approximation, such systems can only provide suboptimal solutions to the problem of joint estimation of time delay and Doppler frequency.
Another prior art technique involves determining values of the following correlation integral
R
x
y
(
τ
,
ω
D
;
ϕ
)
=
1
T
0
∫
0
T
0
x
(
t
-
τ
)
cos
(
ω
D
t
+
ϕ
)
y
(
t
)
ⅆ
t
for the entire interval of hypothesized time delays
&tgr;
min
<&tgr;<&tgr;
max
,
and also for the entire interval of hypothesized Doppler frequencies
&ohgr;
Dmin
<&ohgr;
D
<&ohgr;
Dmax
;
the unknown phase &phgr; should also be varied over the interval (0, 2&pgr;). In this case, values of the correlation integral, determined for some prescribed range of argument values (&tgr;,&ohgr;
D
), define a two-dimensional correlation function. The specific values of arguments &tgr; and &ohgr;
D
, say &tgr;
0
and &ohgr;
D0
, that maximise the two-dimensional correlation function provide estimates of unknown time delay and unknown Doppler frequency.
SUMMARY OF THE INVENTION
FIG. 6
is a block diagram of a suitable system capable of determining values of the correlation integral for some specified time delays &tgr; and Doppler frequencies &ohgr;
D
. The system is formed by a correlator like that of
FIG. 4
, including variable delay line
11
, multiplier
12
and integrator
13
. However, the signal x(t) is first applied to a multiplier
20
which multiplies it by a variable Doppler signal cos(&ohgr;
D
t+&phgr;).
The procedure of adjusting the unknown value of phase &phgr; can be avoided
Alsomiri Isam
Birch & Stewart Kolasch & Birch, LLP
Mitsubishi Denki & Kabushiki Kaisha
Tarcza Thomas H.
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