Pulse or digital communications – Systems using alternating or pulsating current – Plural channels for transmission of a single pulse train
Reexamination Certificate
1999-09-10
2003-05-06
Chin, Stephen (Department: 2634)
Pulse or digital communications
Systems using alternating or pulsating current
Plural channels for transmission of a single pulse train
C348S725000, C375S326000
Reexamination Certificate
active
06560294
ABSTRACT:
STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT
Not applicable.
BACKGROUND OF THE INVENTION
This invention is in the field of data communications, and is more specifically directed to carrier recovery in phase-modulated signals used in such communication.
As is well known in the field, several modulation techniques are now commonly used for the communication of digital signals at high data rates. In general, these modulation techniques are used to encode the communicated digital information into an analog signal by grouping a specified number of bits into a “symbol”, and by then modulating a carrier signal according to the digital value of each symbol in the communicated sequence.
One important way in which such modulation is implemented is phase modulation, in which the value of each communicated signal is at least partially encoded by the phase of the symbol relative to the preceding symbol in the serial stream. In phase-shift keying, for example, the relative inter-symbol phase shift fully specifies the symbol value, and as such no amplitude modulation is involved. A common type of phase-shift keying is quadrature-phase-shift-keying (QPSK), where the relative phase shift from symbol-to-symbol is in multiples of 90° (hence the quadrature nomenclature). QPSK modulation thus encodes two-bit digital symbol values.
Another type of modulation includes both phase and amplitude modulation, such that each symbol is encoded as the combination of an amplitude value (i.e, one value of a defined set of values) and a relative phase shift (also selected as one of a possible set of defined phase shifts). One type of modulation in this class is referred to as quadrature-amplitude-modulation (QAM). QAM modulation techniques are generally referred to by the number of members in their specified “constellation” of values. For example, 16-QAM refers to a modulation scheme in which the symbol amplitude may occupy one of sixteen possible points in complex space. In modern communication systems, 256-QAM has now become popular for encoding eight-bit digital symbol values (i.e., bytes) into 256 possible points in complex space.
In each of these types of modulation, of course, the modulated signals are communicated at a carrier frequency. The carrier frequency determines the rate at which the digital signal values (symbols) are communicated over the particular physical and logical communication facility, whether implemented by coaxial cable, fiber optics, or twisted-pair wires.
Carrier recovery refers to the processes performed at the receiving end of a modulated signal, by way of which the carrier frequency is eliminated from the incoming signal, and the remaining amplitude and phase information is then rendered available for decoding into the digital values for each symbol. In conventional high-performance digital communications receivers, for example cable modems and the like, this carrier recovery is performed in several stages. Typically a first demodulation operation, also referred to as down-mixing, is performed to reduce the incoming signal into, ideally, a spectrum centered about DC. Realistically, the demodulated signal at this point is a substantially low frequency signal that may be represented as follows:
S
(
t
)
e
j[2&pgr;&Dgr;f
0
t+&thgr;
]
+n
where S(t) corresponds to the exact constellation symbols (and thus is a complex quantity, as including phase information). Because such first stage demodulation is not exactly accurate, however, the demodulated signal generally retains a slight phase error that varies over time. In this representation, &thgr;
0
is a phase error for a given symbol, and &Dgr;f
0
is an error frequency at which this phase error varies over time. The n term refers to random noise present in the input signal. In terms of the complex constellation of possible symbol values, one may consider the first-stage demodulated signal as having a phase error corresponding to a rotation of angle &thgr;
0
of the constellation S(t) from its true position, where the rotation varies over time at error frequency &Dgr;f
0
.
Carrier recovery thus also includes a process by way of which the phase errors are eliminated from the demodulated signal, leaving the true complex signal S(t) for decoding. This additional process is often referred to as derotation. A phase-locked loop (PLL) is a commonly used circuit for executing such carrier recovery. As is fundamental in the art, PLLs generally include a phase detection circuit that compares an input signal against the PLL output signal, and that generates an error signal corresponding to the phase difference therebetween; this error signal (typically with high frequency variations filtered out) is then used in modulating the output signal according to the error signal, so that the output signal eventually “locks” onto the input signal. The stable output signal, over time, has time-dependent phase error eliminated therefrom, and is thus suitable for decoding.
In modern high-data rate carrier recovery schemes, it has been observed that the phase detection process is of significant importance. One can increase the data rate of a modulated signal by encoding more bits per symbol, thus increasing the number of points in the modulation constellation. This, of course, also results in smaller phase separation between adjacent constellation points, which necessitates accurate phase detection in the carrier recovery processes. Additionally, the gain of the phase correction produced by the phase detector as a function of phase error is also important, not only in providing high-performance carrier recovery, but also in avoiding false lock situations.
One type of conventional phase estimator is referred to in the art as “power-type estimators”. Attention in this regard is directed to Lindsey and Simon,
Telecommunication Systems Engineering
(General Publishing Company, 1973), pp. 71-80. In these systems, the input signal is raised to a significantly high enough power such that phase information is effectively removed, leaving only information concerning phase error. These power-type phase estimators are useful in pure phase-modulated signals (PSK), but are not particularly suited for modulation schemes, such as QAM, in which the possible phases of the data are not evenly distributed, and in which the phase error cannot therefore be readily retrieved.
Another type of conventional phase detection scheme will now be described relative to
FIG. 1
, in which an example of a conventional carrier recovery circuit is shown. In this example, carrier recovery circuit
2
receives a demodulated input signal of the form:
x′=xe
j&thgr;
+n
where x corresponds to the actual signal, where n corresponds to random (Gaussian white) noise, and where &thgr; is the residual phase error to be removed by carrier recovery circuit
2
. This input signal is applied to one input of multiplier
4
, which applies a phase correction factor e
−j{circumflex over (&thgr;)}
. Low-pass loop filter
8
may include some type of summing or integration, particularly in those cases where phase detector
6
generates a phase estimate {circumflex over (&thgr;)} in the form of a derivative of a probability function.
According to this conventional phase estimation approach, phase detector
6
operates by effectively maximizing a probability function p(&thgr;|x′), and identifying the angle &thgr; that renders this maximum may be considered to be the detected phase error of the input signal x′. After application of Bayes' Rule, and considering both that the phase angle is independent of the constellation point x and also that the probability distribution of phase error &thgr; is uniform, one may consider the following probability function expression:
log
p
(
x
′
|
x
,
θ
)
=
K
-
1
σ
n
2
&LeftDoubleBracketingBar;
x
′
-
x
ⅇ
j
θ
&RightDoubleBracketingBar;
2
where &sgr;
n
2
is the noise power of the Gaussian noise, and where
Brady III W. James
Hernandez Pedro P.
Odom Curtis
Telecky , Jr. Frederick J.
Texas Instruments Incorporated
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