Electricity: motive power systems – Induction motor systems
Reexamination Certificate
2002-04-17
2004-08-17
Le, Dang (Department: 2834)
Electricity: motive power systems
Induction motor systems
C318S814000, C318S801000
Reexamination Certificate
active
06777906
ABSTRACT:
TECHNICAL FIELD
The present invention relates to a control device that controls torque generation of an induction motor with high precision.
BACKGROUND ART
Conventionally, a vector control method is used in induction motors to drive the induction motors with high precision. The vector control method is a method of controlling a d-axis current and a q-axis current on the rotation coordinate axes (d-q axes) that rotate in synchronism with a secondary magnetic flux to respectively desired values. Generally, it is impossible to directly observe the secondary magnetic flux due to hardware limitations. Therefore, there is proposed a sliding frequency vector controlling method in which secondary magnetic flux is estimated based upon a primary current of the induction motor.
However, in this sliding frequency vector controlling method, a value of a secondary resistance is required for calculating the secondary magnetic flux. Therefore, there arises a problem that, changes in the secondary resistance due to heat generation, etc. would cause degradation in the controlling performance.
FIG. 12
is a graph which is obtained by plotting the relationship between the torque instruction and the torque error in a conventional control device for an induction motor to which the sliding frequency vector controlling system is applied. In
FIG. 12
, the abscissa represents the torque instruction, and the ordinate represents the torque error (=generated torque-torque instruction). The graph in the upper half of the
FIG. 12
represents the relationship between the torque instruction and the torque error when the rotation speed is 3 [rad/s], and the graph in the lower half represents the relationship between the torque instruction and the torque error when the rotation speed is 188 [rad/s]. Moreover, the solid line shows the characteristic obtained when the secondary resistance of the induction motor has increased by a factor of 1.3, and the broken line shows the characteristic obtained when the secondary resistance of the induction motor has decreased by a factor of {fraction (1/1.3)}.
As shown in
FIG. 12
, in the conventional control device for the induction motor to which the sliding frequency vector controlling system has been applied, there arises a problem that, when the value of the secondary resistance changes, a torque error is generated irrespective of the rotation speed.
A method which identifies a secondary resistance value during the driving operation of the induction motor has been proposed. However, depending on driving conditions, the estimated value of the secondary resistance is dispersed, resulting in problems with respect to the stability.
In order to solve these problems, a control device for an induction motor including a magnetic flux observing device, which calculates the secondary magnetic flux based upon the primary current and the primary voltage of the induction motor, and to which the induction motor constant is applied, has been proposed.
For example,
FIG. 13
is a structural diagram which shows a prior art control device for an induction motor shown in a document “Indirect Field Oriented Control Method Using Flux Observer Equivalent To The Direct Field Oriented Control Method” (1992 National Convention Record of the IEE Japan-Industry Application Society, No.110 (pp. 466-471).
The principle of controlling by the prior art control device for an induction motor will be explained. The magnetic flux observing device, which is constituted on stationary bi-axes (&agr;-&bgr; axes), is designed based upon the following equations (1) and (2).
ⅆ
ⅆ
t
(
i
^
α
s
i
^
β
s
)
=
A
11
(
i
^
α
s
i
^
β
s
)
+
A
12
(
φ
^
α
r
φ
^
β
r
)
+
B
(
v
α
s
v
β
s
)
+
K
1
(
i
^
α
s
-
i
α
s
i
^
β
s
-
i
β
s
)
(
1
)
ⅆ
ⅆ
t
(
φ
^
α
r
φ
^
β
r
)
=
A
21
(
i
^
α
s
i
^
β
s
)
+
A
22
(
φ
^
α
r
φ
^
β
r
)
+
K
2
(
i
^
α
s
-
i
α
s
i
^
β
s
-
i
β
s
)
(
2
)
where,
A
11
=
(
-
(
R
s
σ
L
s
+
R
r
(
1
-
σ
)
σ
L
r
)
0
0
-
(
R
s
σ
L
s
+
R
r
(
1
-
σ
)
σ
L
r
)
)
A
12
=
(
MR
r
σ
L
s
L
r
2
P
m
ω
m
M
σ
L
s
L
r
-
P
m
ω
m
M
σ
L
s
L
r
MR
r
σ
L
s
L
r
2
)
A
21
=
(
MR
r
L
r
0
0
MR
r
L
r
)
A
22
=
(
-
R
r
L
r
-
P
m
ω
m
P
m
ω
m
-
R
r
L
r
)
B
=
(
1
σ
L
s
0
0
1
σ
L
s
)
In order to place the pole of the magnetic flux observing device on a conjugate complex pole or a duplex pole, square matrixes K
1
, K
2
are defined by using equations (3) and (4), and k
1
, k
2
, k
3
, k
4
are determined according to the rotation speed.
K
1
=
(
k
1
-
k
2
k
2
k
1
)
(
3
)
K
2
=
(
k
3
-
k
4
k
4
k
3
)
(
4
)
Therefore, when equation (1) is coordinate-converted onto rotation axes (d-q axes) with equation (2) being coordinate-converted onto the stator polar coordinates, equations (5) to (7) are obtained.
ⅆ
ⅆ
t
(
i
^
d
s
i
^
q
s
)
=
A
~
11
(
i
^
d
s
i
^
q
s
)
+
(
MR
r
σ
L
s
L
r
2
-
P
m
ω
m
M
σ
L
s
L
r
)
φ
^
ds
+
B
(
v
ds
v
qs
)
-
K
1
(
i
^
d
s
-
i
ds
i
^
q
s
-
i
qs
)
(
5
)
ⅆ
ⅆ
t
φ
^
dr
=
-
R
r
L
r
φ
^
dr
+
MR
r
L
r
i
^
ds
-
(
k
3
(
i
^
ds
-
i
ds
)
-
k
4
(
i
^
qs
-
i
qs
)
)
(
6
)
ⅆ
ⅆ
t
θ
^
(
=
ω
)
=
P
m
ω
m
+
MR
r
L
r
i
^
qs
φ
^
dr
-
k
4
(
i
^
ds
-
i
ds
)
+
k
3
(
i
^
qs
-
i
qs
)
φ
^
dr
(
7
)
where the following equation is satisfied.
A
~
11
=
A
11
-
(
0
-
ω
ω
0
)
In other words, based upon equations (5) to (7), magnetic flux calculations that are equivalent to the magnetic flux observing device on the &agr;-&bgr; axes can be obtained on the d-q axes.
With respect to the two square matrixes K
1
, K
2
, those that have been designed on the stationary bi-axes are applied. In other words, square matrixes K
1
, K
2
are defined by equation (3) and equation (4), and according to rotation speeds, k
1
, k
21
, k
3
and k
4
are determined. In this case, between K
1
and K2, the relationship represented by equation (8) is always satisfied.
K
1
K
2
=K
2
K
1
(8)
Here, the relationship between mutually commutative matrixes K
1
, K
2
refers to the relationship satisfying a commutative rule. Here, equations (5) to (7) can be rewritten to equations (9) to (13).
(
z
1
z
2
)
=
K
1
(
i
^
ds
-
i
ds
i
^
qs
-
i
qs
)
(
9
)
(
z
3
z
4
)
=
K
2
(
i
^
ds
-
i
ds
i
^
qs
-
i
qs
)
(
10
)
ⅆ
ⅆ
t
(
i
^
ds
i
^
qs
)
=
A
~
11
(
i
^
ds
i
^
qs
)
+
(
MR
r
σ
L
s
L
r
2
-
P
m
ω
m
M
σ
L
s
L
r
)
φ
^
ds
+
B
(
v
ds
v
qs
)
-
(
z
1
z
2
)
(
11
)
ⅆ
ⅆ
t
φ
^
dr
=
-
R
r
L
r
φ
^
dr
+
MR
r
L
r
i
^
ds
-
z
3
(
12
)
ⅆ
ⅆ
t
θ
^
(
=
ω
)
=
P
m
ω
m
+
(
MR
r
L
r
i
^
qs
-
z
4
)
+
φ
^
dr
(
13
)
Therefore, based upon equations (9) to (13), it is possible to obtain the phase and amplitude of an estimated secondary magnetic flux with the same precision as a magnetic flux observing
Kinpara Yoshihiko
Sakurai Hisao
Terada Kei
Le Dang
Leydig , Voit & Mayer, Ltd.
Mitsubishi Denki & Kabushiki Kaisha
Scheuermann David W.
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