Electricity: power supply or regulation systems – In shunt with source or load – Using a three or more terminal semiconductive device
Reexamination Certificate
1999-02-01
2001-05-08
Wong, Peter S. (Department: 2838)
Electricity: power supply or regulation systems
In shunt with source or load
Using a three or more terminal semiconductive device
C323S283000
Reexamination Certificate
active
06229288
ABSTRACT:
TECHNICAL FIELD
The invention relates to the field of electrical power conversion and control, across a wide range of current and voltages; more particularly, it relates to method and apparatus for electronically implemented power control, more particularly, those implemented with switch mode power conversion techniques.
BACKGROUND OF THE INVENTION
An electrical power conversion circuit is a circuit in which electrical power is changed so that a power source with a voltage or current can serve a load requiring a different voltage or current. In switch mode conversion circuits and power conversion techniques, power is typically changed from a supply that is higher in current or voltage to serve a load requiring lower current or voltage, or from a supply that is lower in voltage or current to a load requiring higher voltage or current. Where high conversion efficiency is made possible by design of the circuit, power output is substantially equal to power input. Where voltage-ampere products (VA) are close to the same at both output and input, and assuming a fixed VA at the input, a reduction in voltage at a resistive load is necessarily accompanied by an increase in load current, and vice-versa.
There are three basic topologies of switch mode power converters. They include step-down (buck—see FIG.
1
), step-up (boost), inverting (including flyback) converters, and “duals” of these three. A “dual” version of any of the three basic class devices can sometimes be effected by a simple transform as follows: series inductors become parallel capacitors; and parallel capacitors becomes series inductors. Other transforms can be more involved, and the above transforms are provided as illustrative only, as will be appreciated by those skilled in the art. Dual topologies are known to have the following characteristics: a) discontinuous currents to and from the voltage sources become continuous currents to and from voltage sources; b) the DC transfer function (output voltage vs. input voltage vs. duty cycle) remains the same; and c) the dual input and output inductors can be combined together in one magnetic structure.
In all such topologies, at least with respect to DC functionality, each converter typically consists of two switches, an inductor, and input and output filters. Nearly all conventional converters are some derivation or combination of these topologies and their duals.
Output voltage regulation in these known converter topologies then is achieved by varying the duty cycle of the switches.
FIG. 1
shows a simple buck converter with ideal switches. In DC conversion circuits of the type generally known as buck or boost regulators, the two solid state switches typically employed are reciprocally and cyclically operated so that one switch is “on” or conducting, while the other is “off” or non-conducting, and vice-versa. Thus as the duty cycle of the modulation of the two switches is varied, so is the voltage (or current) conversion ratio varied between source and load. For example, in a buck topology, if S
1
is modulated at a given duty cycle D and S
2
is modulated exactly opposite S
1
(S
2
closed when S
1
is open and vice versa) then output voltage is given by the formula:
V
out
=V
in
*D.
Other known formulae similarly apply to other respective known regulator classes, as will be appreciated by those skilled in the art. This relationship holds for either polarity of V
in
. Theoretically then, alternating voltage on the input would manifest itself on the output according to the same relationship, assuming the use of “ideal” switches. However, as a matter of practice, in the absence of such ideal switches, conventional single stage converter implementations do not function in AC to AC conversions.
In conversion circuits used to drive reactive loads such as induction motors, bi-directional energy flows and other four quadrant operation must also be accommodated. Simple power transformers are in common usage, though necessarily restricted to AC power conversion; however, they grow heavy and bulky as power levels increase, and they are by nature not readily variable in their conversion ratios without some kind of tap changing modification.
There has been substantial work done in the area of AC converters, but known methods suffer for one or more reasons. For instance, some employ simplistic control schemes that lead to a variety of failure modes in the switches.
In all real world switches, there exist timing delays and finite rise/fall times, both of which vary from device to device and over varying operating conditions. If care is not taken in a conventional two switch converter as outlined above, both switches could conduct simultaneously, with attendant high currents and excessive power dissipation which can destroy the switches.
Power transistors of some of the types commonly employed as switches in converter topologies (and other semiconductors similarly employed) are known to store significant amounts of charge, and if a control voltage is applied to turn one transistor off as the control voltage is being applied to turn the other transistor on, the flow of current in the first transistor would continue for sometime after the turn off control, and simultaneous conduction in both transistors would occur to cause a short across the power source, with potentially damaging current flow through the switches.
A simultaneous “off” condition for both transistors is also a problem, for if the first transistor is turned off before the second transistor is turned on, the series inductor in such regulating circuits (in series with the opening switch) would discharge, or force current, through the opening switch and subject the switch to potentially damaging voltage.
One known technique for dealing with the first phenomenon is the addition of a switching delay, or dead time, into the turn-on of each switch, after turn-off of the other switch. Generally, a value for the length of the switching delay is chosen to insure that one switch is completely off before the other is enabled. But in the AC circuit supposed above, that then results in both switches being disabled at the same time. And as discussed above, if any current is flowing in the output inductor L
1
, then the result of both switches simultaneously disabled is a voltage spike across the switches which will likely destroy them. This spike typically has to be clamped via some snubber or clamping network, but that then results in excessive clamp power dissipation and excessive switching losses in the switches. For an example of a manifestation of this problem, and an example of this limiting solution, see U.S. Pat. No. 4,947,311 to Peterson, the disclosure of which is hereby incorporated by this reference into this disclosure as background as if fully set forth.
Another approach to the problems described above has been through the use of resonant switching circuits that employ zero voltage or current switching techniques. These are sometimes referred to as “soft switching”, or zero voltage switched, techniques. Converters employing these techniques do tend to be more efficient in theory than the “hard” switching circuits using snubbers mentioned above, but some topologies have proven difficult to control, where the resonant circuit becomes increasingly unstable at lower power ranges. In addition, such resonant circuits also have more narrowly defined operating conditions (i.e., minimum and maximum current limitations), and are therefore less robust for industrial applications, and these circuits typically trade switch losses for increased conduction losses, and require bulky resonant componentry.
Switching losses into an inductive load, as encountered in conventional circuits, are generally proportional to the product of turnoff time, peak current, peak voltage, and switching frequency, and can be calculated from the well known formula:
P
sw
=0.5
t
off
*V
peak
*I
peak
*Frequency
where P
sw
is the switching loss expressed in units of power. In a snubbed or clamped circuit, there is always a voltage rise across the
Baretich David F.
Wiegand Gregory P.
Dwyer Patrick M.
MicroPlanet Ltd.
Vu Bao Q.
Wong Peter S.
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