Low voltage current sense amplifier circuit

Details Low voltage current sense amplifier circuit Low voltage current sense amplifier circuit

2001-12-27

2002-12-10

Tran, Toan (Department: 2816)

Miscellaneous active electrical nonlinear devices, circuits, and

Specific signal discriminating without subsequent control

By amplitude

C327S066000, C327S561000

Reexamination Certificate

active

06492845

ABSTRACT:

BACKGROUND OF THE INVENTION
1. Technical Field of the Invention
The present invention relates to the field of current sense amplifiers.
2. Description of Related Art
Current sense amplifiers are typically used to measure the amount of current supplied by and to a device or component in various types of electronic equipment. Reference is now made to
FIG. 1
wherein there is shown a schematic diagram of a typical implementation for a current sense amplifier
10
. A current (referred to as I
load
) to be sensed flows through a sense resistor (R
sense
) from a first sensing node (R
s+
) to a second sensing node (R
s−
). The first sensing node (R
s+
) is connected to a first input (+) of an operational amplifier
12
through a resistor R
1
. The second sensing node (R
s−
) is connected to a second input (−) of the operational amplifier
12
through a resistor R
2
. The output of the operational amplifier
12
is connected to the base of a transistor
14
whose collector is connected to the first input (+). A current (referred to as I
ref
) flows through the connection between the first input (+) and the collector of the transistor
14
. An emitter of the transmitter
14
, generates an input current (referred to as I
in
) and, is connected to an input of a 1:n current mirror
16
. Vcc is connected to an output of the current mirror
16
through a load resistor R
3
for the generation of an output current (referred to as I
out
) The second input (−) of the operational amplifier
12
presents a relatively high impedance. Neglecting the presence of any input bias current at the second input (−), it is recognized that no current flows through the second resistor R
2
from the second sensing node (R
s−
) . The voltage at both the first input (+) and second input (−) is therefore equal to the voltage at the second sensing node (R
s−
). The voltage drop across the resistor R
1
is accordingly equal to the product of I
load
and R
sense
. The input current I
in
thus equals the current I
ref
, and ideally then:
I
ref
=I
load
·(
R
sense
/R
1
);  (1)
and
I
out
=n·I
in
;  (2)
and
I
out
=n·I
load
·(
R
sense
/R
1
).  (3)
In practice, however, the value of base current at the transistor
14
cannot be neglected as it is also multiplied by a factor of n in the current mirror
16
and alters the value of the output current I
out
(of Equation 3) away from ideal. Still further, it is recognized that a current mirror
16
possessing a large factor (for example, n equals approximately twenty for a Wilson current mirror) is not particularly accurate. The value of the input current is actually set as follows:
I
in
=I
ref
·(1+1/&bgr;)  (4)
wherein: &bgr; is the current gain of the transistor
14
.
Reference is now made to
FIG. 2
wherein there is shown a schematic diagram of the 1:n current mirror
16
. This current mirror has a conventional configuration that is well known to those skilled in the art. A detailed description of the components, interconnection and operation of the current mirror
16
is accordingly not required. Continuing with the foregoing analysis, and specifically with respect to the current mirror
16
, the relationship between the input current I
in
and the output current I
out
is given by the following:
I
out
=
I
in
·
n
·
β
2
+
n
·
β
+
β
β
2
+
(
n
+
1
)
·
β
+
n
+
1
(
5
)
wherein: &bgr; is the current gain of the matched transistors within the current mirror
16
. From Equations 4 and 5, the actual value of the output current I
out
is given by the following:
I
out
=
n
·
I
ref
·
(
1
-
(
n
-
1
n
-
1
)
·
β
+
(
n
-
1
n
)
β
2
+
(
n
+
1
)
·
β
+
n
+
1
)
;
 
⁢
and
(
6
)
by taking into account Equation (1);
I
out
=
n
·
I
load
·
(
R
sense
R1
)
·
(
1
-
(
n
-
1
n
-
1
)
·
β
+
(
n
-
1
n
)
β
2
+
(
n
+
1
)
·
β
+
n
+
1
)
.
(
7
)
Now, from a comparison of the foregoing Equations, it is recognized that the actual output current (see, Equation 7) of the current mirror
16
and the current n·I
ref
that should preferably (and ideally) be output from the current mirror approximately differ from each other by a factor (shown inside the parenthetical of Equation 7) on the order of:
(
n
-
1
n
-
1
)
/
β
.
Given a scenario where n is relatively large (for example, greater than or about ten) and &bgr; is relatively small (for example, less than or about sixty), this factor can present a significant difference in measured current. In this configuration, the current sense amplifier circuit of
FIG. 1
cannot be used for generating a precision current amplifier output.
SUMMARY OF THE INVENTION
A transconductance amplifier measures a current passing through a sense resistor to generate a reference current indicative of the measured current. A current mirror circuit connected to the transconductance amplifier amplifies the reference current to generate an amplified output current. A cascode circuit is connected between the current mirror circuit and output of the generated amplified output current.


REFERENCES:
patent: 5627494 (1997-05-01), Somerville
patent: 5923217 (1999-07-01), Durec
patent: 5969574 (1999-10-01), Legates
patent: 6011415 (2000-01-01), Hahn et al.
patent: 6049469 (2000-04-01), Forbes et al.
patent: 6392392 (2002-05-01), Nakahara
Maxim Integrated Products, 19-1184: Rev. 0: Dec. 1996, Maxim Low-Cost, Precision, High-Side Current-Sense Amplifier, MAX4172, pp. 1, 6.

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