Telecommunications – Transmitter – Having harmonic radiation suppression
Reexamination Certificate
2000-05-03
2003-11-25
Maung, Nay (Department: 2745)
Telecommunications
Transmitter
Having harmonic radiation suppression
C455S127500, C375S297000
Reexamination Certificate
active
06654591
ABSTRACT:
BACKGROUND OF THE INVENTION
1. Field of the Invention
This invention relates to a signal amplification system and, more particularly, to a system and method which enables linear amplification of a signal.
2. Description of Related Art
An ideal power amplifier amplifies an input signal with no waveshape alteration. The ideal power amplifier is therefore characterized as having a transfer function (input signal vs. output signal) which is linear with no transfer function discontinuities. In practice, however, a power amplifier has a transfer function with nonlinear and “linear” regions. Whether the power amplifier is operating in a linear or nonlinear region depends on the amplitude of the input signal. For the power amplifier to achieve as near to linear operation as possible, the power amplifier is designed to operate within its linear region given the range of possible input signal amplitudes. If the input signal has an amplitude which causes the power amplifier to operate outside the linear region, the power amplifier introduces nonlinear components or distortion to the signal. When the input signal possesses peak amplitudes which cause the amplifier to compress, to saturate (no appreciable increase in output amplitude with an increase in input amplitude) or to shut-off (no appreciable decrease in output amplitude with a decrease in input amplitude), the amplifier is being overdriven, and the output signal is clipped or distorted in a nonlinear fashion. Generally, an amplifier is characterized as having a clipping threshold, and input signals having amplitudes beyond the clipping threshold are clipped at the amplifier output. In addition to distorting the signal, the clipping or nonlinear distortion of the input signal generates spectral regrowth or adjacent channel power (ACP) that can interfere with an adjacent frequency.
Various linearization methods are used to enable the use of more cost-effective and more power efficient amplifiers while maintaining an acceptable level of linearity. Feed-forward correction is routinely deployed in modern amplifiers to improve the linearity of the main amplifier with various input patterns. The essence of the feed-forward correction is to sample the main amplifier output, isolate the distortion components generated from the main amplifier on a feed forward path by canceling the main signal components from the feed forward path. The distortion components are provided to a linear correction amplifier on the feed forward path which amplifies the distortion components. The distortion components on the feed forward path are maintained at 180 degrees out of phase to the distortion components on the main signal path and are combined with the distortion components on the main signal path. As the combined distortion components are 180 degrees out of phase, the distortion components cancel without affecting the main signal, thus providing a linear signal at the feed forward amplifier output.
Another linearization technique involves splitting a signal to be amplified by separate amplifiers of the same gain and power performance, and the amplified signal components are constructively combined at the output while the distortion components are used to cancel each other.
FIG. 1
shows an amplifier circuit for amplifying an input signal S
in
to produce an amplified output signal S
out
. The input signal S
in
can include CDMA or TDMA modulated RF carrier signals having respective fundamental frequencies f
1
and f
2
. Both frequencies or signal components f
1
and f
2
can lie within standard wireless frequency bands in the 800-960 MHz vicinity. The various signals are shown in a vectorial fashion to conveniently illustrate phase relationships between the same frequency components at various points within the circuit
10
. Thus a vector pointing in an upwards direction represents a frequency component of the opposite phase as the same frequency component represented by a downwardly pointing vector.
Input signal S
in
is applied to input port
12
of a first coupler or power splitter
14
which splits signal into signal S
1
at a coupled path output port
16
and a signal S
2
at a direct path output port
18
. The coupler
14
is preferably a passive device which may be a conventional branch line coupler or Wilkinson type divider that splits input power unequally between the two output ports, preferably with higher power being provided at port
18
. For example, the signal level of signal S
2
may be 10 dB higher than that of the signal S
1
. In this embodiment, the frequency signal components f
1
and f
2
produced on the direct path port
18
are delayed by a 90 degree phase shift while the frequency signal components on the coupled path port
16
have a 0 degree phase shift or no phase shift. Signal S
1
contains only the frequency signal components f
1
and f
2
and is applied to a first amplifier
20
(A
1
) where it is amplified to produce an amplified signal S
3
at the amplifier output. The amplifier
20
(A
1
) can be a conventional high frequency amplifier operating in class A, AB or B with power gain on the order of 30 dB to produce RF output power of 50 Watts, for example.
As is well known in the art, when a dual or multi-tone signal is applied to an amplifier, which is not perfectly linear, IMD products are generated at predictable frequencies. These IMD products are particularly apparent when the amplifier is being operated in saturation or in the gain compression region of the amplifier. The further into the gain compression region the amplifier is operated, the higher will be the IMD product levels. In addition, amplifiers which operate in class AB or class B modes tend to produce high IMD product levels when multi-frequency input signals are amplified. IMD product levels on the order of −30 dBc (30 decibels below the fundamental frequency or carrier level) are typical. Amplified signal S
3
contains amplified frequency signal components f
1
and f
2
as well as undesirable intermodulation distortion (IMD) products or distortion components at frequencies f
3
and f
4
, where f
3
is typically a lower frequency than f
1
and f
4
is a higher frequency than f
2
. The frequency signal components f
1
and f
2
of the signal S
3
are designated as having a zero degree phase shift, and the distortion components f
3
and f
4
are also designated as having a zero degree phase shift in brackets.
The amplified signal S
3
is applied to input port
22
of coupler
24
, which may be a conventional hybrid (e.g. branch line), backward firing or parallel-line coupler with a coupling value C
22
. In this case, coupled path signal S
4
on output coupling port
26
will be 30 dB below the level of the direct path signal S
8
emanating from direct port
25
. The voltage levels of the frequency components of the signal S
4
are each C
22
times the corresponding voltage levels of the S
3
frequency components. The voltage levels of the S
8
are the square root of {square root over (1−C
22
2
)} times the corresponding voltage levels of the S
3
frequency components. The phases of the frequency components of S
4
will be equal to the corresponding ones of S
8
. When a branch line or other hybrid coupler is used for the coupler
24
, then the signals S
4
and S
8
will differ in phase by 90 degrees. In this embodiment, the coupler
24
produces the signal S
8
from the direct port
25
with the frequency components f
1
-f
4
phase shifted by 90 degrees as designated by the −90 degrees and the −90 degrees in brackets. The phases of the frequency components f
1
-f
4
of coupled path signal S
4
from the coupling port
26
remain at 0 degrees.
Coupled path signal S
4
is then applied to an attenuator
27
and a phase shifter
28
. The attenuator
27
and the phase shifter
28
are designed to adjust the amplitude and phase of the signal S
4
at each of the frequencies f
1
-f
4
. The amplitude of the frequency signal components f
1
and f
2
of the resulting signal S
5
is adjusted to be smaller in amplitude than the frequency signal co
Garceran Julio A.
Lucent Technologies - Inc.
Maung Nay
Trinh Tan
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