Telecommunications – Transmitter – Amplitude modulation
Reexamination Certificate
2001-05-29
2003-12-16
Vuong, Quochien (Department: 2685)
Telecommunications
Transmitter
Amplitude modulation
C455S127200, C375S300000
Reexamination Certificate
active
06665525
ABSTRACT:
BACKGROUND OF THE INVENTION
The present invention relates to amplitude modulation and more particularly to amplitude modulation for radio transmitters.
When a transmitter power amplifier must faithfully amplify a signal of varying amplitude and phase, such as a single sideband voice signal, or a digitally modulated signal, such as 16 Quadrature Amplitude Modulation (16QAM) or linear 8-level Phase Shift Keying (8-PSK), a linear amplifier has most often been used in the prior art. Linear amplifiers are typically of lower efficiency than saturated, constant envelope amplifiers, and are not perfectly linear, giving rise to intermodulation distortion. As such, the prior art has attempted various improvements to linear amplification techniques aimed at improving efficiency or linearity.
An arbitrarily modulated signal can also be amplified by using a non-linear, e.g. saturated, power amplifier to amplify a drive signal modulated with the varying phase of the desired signal while amplitude modulating the power amplifier with the varying amplitude of the desired signal. Conventionally, the amplitude modulation could include high-level amplitude modulation in which the power supply voltage to the amplifier is modulated, including the use of a pulse-width modulated power supply to modulate the voltage.
Such conventional high-level amplitude modulation, however, may be limited in its ability to modulate the power amplifier over a wide dynamic range of desired amplitudes or output power levels, and may also exhibit some form of distortion when the load impedance deviates from an ideal match. Conventionally, an isolator has been used to isolate the power amplifier from the load impedance mismatch. However, isolators are typically large and expensive components and, therefore, situations may arise where it is impractical to use an isolator.
FIG. 1A
shows a conventional power amplifier that is high-level amplitude modulated by controlling its supply voltage. A representation of the desired amplitude between zero and 100% may be provided by, for example, digital signal processing. For example, the digital signal processing can generate a Sigma-Delta representation of the desired amplitude modulation waveform in which the instantaneous modulation level between zero and 100% is represented by the proportion of binary “1”s in a digital bitstream. Generally, such a representation has the advantage that conversion to an analog waveform requires merely low-pass filtering. Thus,
FIG. 1A
shows a sigma-delta amplitude waveform entering the input of level-shifter
20
, which has the function of scaling the digital signal so that a “1” is represented by the maximum power amplifier supply voltage “V
battery
” while a binary “0” is represented by a zero voltage, or the other pole of the supply, if not zero voltage. The scaled sigma-delta waveform is now low-pass filtered using a filter
21
which has a bandwidth wide enough to pass all significant amplitude modulation components while attenuating the sigma-delta quantizing noise. Sigma delta converters may be of the higher order type (e.g. order 2 or 3) to suppress quantizing noise that falls within the passband width of the filter
21
.
The filtered amplitude modulated (AM) representation from the filter
21
comprises a voltage waveform that instantaneously lies between zero and V
battery
and undergoes excursions between these limits. The actual supply voltage on the power amplifier
24
is compared by the comparator
22
with the filtered AM waveform. If the supply voltage is lower than the AM voltage then the comparator
22
changes the control electrode voltage on series regulating transistor
13
so as to increase the supply voltage, and vice versa, thereby controlling the voltage to the power amplifier (PA)
24
to follow
20
the desired AM waveform. The series regulating transistor
13
may be a P-type field effect transistor constructed in a diffused metal-oxide-semiconductor (DMOS) or VMOS process which gives low on state resistance, thereby typically preventing loss of voltage when the AM signal demands maximum voltage. In the case of a reverse polarity circuit with V
battery
negative relative to ground, an N-type VMOS field effect transistor (FET) could be used.
When the PA
24
is constructed with Gallium Arsenide (GaAs) metal-semiconductor field effect transistor (MESFET) devices, the output signal amplitude delivered to the load typically follows the desired AM waveform applied to the PA supply voltage fairly closely down to small voltages and low signal output levels. However, when GaAs Heterojunction Bipolar Transistors (HBTs) are used for the PA
24
, the output signal amplitude typically does not follow variations in the modulated supply voltage down to low levels. Typically, the output of an HBT amplifier tends to fall more rapidly than the supply voltage at lower levels. However, both MESFET and HBT PAs may tend to exhibit a more linear relationship between output signal amplitude and current consumption. This is demonstrated by the measured data in the graphs of
FIGS. 1B and 1C
which illustrate output RF amplitude as a function of modulated supply voltage (
FIG. 1B
) and as a function of modulated supply current (
FIG. 1C
) for a FET and a HBT power amplifier.
FIG. 2
shows a power amplifier that is high-level amplitude modulated by controlling its supply current rather than its voltage. The level-shifter
20
and the filter
21
produce the same AM waveform as in FIG.
1
A. The comparator
22
compares the instantaneous AM waveform voltage with a voltage signal from current-to-voltage converter
27
, which may include a sense resistor
26
and an operational amplifier
25
, which senses the current flowing through series regulator transistor
13
to the PA
24
by amplifying the voltage drop across current sensing resistor
26
of, for example, 0.1 ohms, utilizing amplifier
25
. The scaling of the current sensor circuit may be determined by resistor
26
and amplifier
25
such that the current range (zero to maximum current) produces an output voltage of zero to V
battery
. In this way, the AM signal from filter
21
, which ranges between 0 and V
battery
, controls the PA current over the corresponding range zero to I
max
. I
max
is the current that flows in the PA
24
when its supply voltage equals V
battery
and the load impedance is nominal (matched). Thus, at least at the two extreme ends of the range (zero to V
battery
for voltage modulation and zero to I
max
for current modulation), with either the voltage control of
FIG. 1A
or the current control of
FIG. 2
, the PA
24
may deliver the same output power and amplitude (at least when the load impedance is nominally correct).
If the load impedance is not correct, for example, if it is half the ideal value, (such as a 2:1 voltage standing wave ratio (VSWR) on the low side) then the voltage control circuit of
FIG. 1A
will generally apply the same supply voltage waveform to the PA
24
as if the load impedance is nominally correct, and the PA
24
will attempt to deliver the same output voltage to the load. However, the load current and the PA current will double when the load impedance is halved, and this might exceed the current delivery capability of the PA
24
. In that case the PA
24
would come out of saturation and the power output would typically limit or clip before the supply voltage had modulated up to 100% of V
battery
, which may cause modulation distortion.
Similarly, if the load impedance is twice the ideal value (a VSWR of 2 on the high side), then the current control circuit of
FIG. 2
will typically control the PA current to be the same as with a nominal load, but the same output current flowing into twice the impedance will cause the load voltage to double. This may exceed the capacity of the PA
24
to deliver voltage to the load, and the output power may limit or clip before the current has been modulated up to 100% of I
max
, which may cause modulation distortion.
SUMMARY OF THE INVENTION
Embodiments of the present invention provide metho
Dent Paul W.
Hadjichristos Aristotle
Pehlke David R.
Ericsson Inc.
Myers Bigel Sibley & Sajovec P.A.
Vuong Quochien
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