Pulse or digital communications – Receivers – Angle modulation
Reexamination Certificate
2001-03-06
2004-01-06
Bayard, Emmanuel (Department: 2631)
Pulse or digital communications
Receivers
Angle modulation
Reexamination Certificate
active
06674814
ABSTRACT:
TECHNICAL FIELD
The present invention relates to a frequency error estimating apparatus for estimating a frequency error or a frequency differential between a local oscillation frequency and a carrier frequency of a received signal in a receiver used for a satellite communication, a mobile satellite communication and a mobile communication. The invention particularly relates to a frequency error estimating apparatus capable of securing high precision in estimating a frequency error without losing the level of following the time variation in the Doppler frequency.
BACKGROUND ART
A conventional frequency error estimating apparatus will be explained below. Some of the receivers used for mobile communications employ a synchronous detection system that is capable of obtaining satisfactory detection characteristics even in a low C/N channel. According to this synchronous detection system, a carrier that is synchronous with a carrier frequency of a received signal is reproduced, and a detection output is obtained based on this carrier. However, the receiver that employs the synchronous detection system has variations in the oscillation frequency due to variations in the precision of the oscillator, variations in temperature, etc. As a result, there arises a difference in the frequency between the transmitter and the receiver. In other words, a frequency error occurs between the transmitter and the receiver. When such a frequency error exists, the phase at a signal point on an IQ plane (a complex plane expressed by a real axis and an imaginary axis) rotates.
Therefore, in order to minimize this frequency error and to improve the synchronous characteristics in the receiver, it becomes necessary to provide a frequency error estimating apparatus that measures a phase rotation volume from a received signal and estimates a frequency error from a result of this measurement.
FIG. 8
shows a structure of a conventional frequency error estimating apparatus. This frequency error estimating apparatus is disclosed in “Doppler-Corrected Differential Detection of MPSK”, IEEE Trans. Commun., Vol. COM-37, 2, pp. 99-109, Feb., 1989. In
FIG. 8
, legend
1
denotes a received signal, legend
21
denotes an M-multiplier for removing a modulation component of the received signal
1
, and legend
22
denotes a D-symbol differential detector for performing differential detection over a period of D symbols based on the output of the M-multiplier
21
. Legend
101
denotes an averaging filter for averaging the output of the D-symbol differential detector
22
, and thereby suppressing a noise component. Legend
24
denotes a coordinate converter for calculating a phase component from the output of the averaging filter
101
, and legend
25
denotes a divider for calculating a frequency error from a phase component that has been output from the coordinate converter
24
. Legend
2
denotes an estimated frequency error value that is output from the divider
25
.
FIG. 9
shows an example of an internal structure of the averaging filter
101
. This shows an IIR (a primary infinite impulse response) filter. In
FIG. 9
, legends
111
and
112
denote multipliers for multiplying an input signal by a specific coefficient respectively. Legend
32
denotes an adder for adding two inputs, and
33
denotes a delay unit for delaying a signal by one symbol.
The conventional frequency error estimating apparatus having the above-described structure is a D-symbol differential detection type frequency error estimating apparatus that estimates a phase change volume due to a frequency error, by performing differential detection over a period of D symbols based on a received signal.
The operation principle of the conventional frequency error estimating apparatus will be explained with reference to FIG.
8
and FIG.
9
. When a modulation system used is the M-phase PSK (phase shift keying) system, the received signal 1 (r(nT)) is expressed by the following equation (1).
r
(
nT
)=
A
(
nT
)exp[
j
{&thgr;(
nT
)+&Dgr;&ohgr;
nT}]
(1)
In the equation (1), the received signal
1
(r(nT)) is a complex base band signal sampled in a symbol period T. A(nT) expresses an amplitude component, and &Dgr;&ohgr; expresses an angular frequency error. &thgr;(nT) expresses a modulation component, and this takes M values of, for example, 2&pgr;k/M (k=0, 1, . . . , and M−1). To simplify the explanation, it is assumed that there is no noise component.
The M-multiplier
21
multiplies the received signal
1
by a modulation multiple number M for removing the modulation component of the received signal
1
. A signal after the multiplication (r1(nT)) is expressed by the following equation (2).
r
1(
nT
)=
A
(
nT
)exp[
jM
{&thgr;(
nT
)+&Dgr;&ohgr;
nT}]
(2)
In the equation (2), M&thgr;(nT) is a multiple of 2&pgr;, and therefore, this can be disregarded. The equation (2) can be substituted by the following equation (3).
r
1(
nT
)=
A
(
nT
)exp(
jM&Dgr;&ohgr;nT
) (3)
The D-symbol differential detector
22
performs differential detection over a period of D symbols based on the output (r1(n)) from the M-multiplier
21
. A signal after the differential detection (d1(nT)) is expressed by the following equation (4).
d
1(
nT
)=
r
1(
nt
)
r
1*(
nT−DT
)
=A
(
nT
)
A
(
nT−DT
)exp(
jMD&Dgr;&ohgr;T
) (4)
In the equation (4), r1* (nT−DT) is a conjugate complex number of r1(nT−DT).
The averaging filter
101
averages the output (d1(nT) of the D-symbol differential detector
22
, and thereby suppresses the noise component. For example, when the primary IIR filter shown in
FIG. 9
is used as the averaging filter, an output (d2(nT)) of the averaging filter
101
is expressed by the following equation (5).
d
2(
nT
)=&agr;
d
1(
nT
)+(1−&agr;)
d
2(
nT−T
) (5)
In the equation (5), the first term is a result of the multiplier
111
multiplying the input signal (d1(nT)) by the coefficient &agr;, and the second term is a result of the multiplier
112
multiplying the one symbol-delayed output (d2(nT−T)) of the averaging filter
101
by the coefficient 1−&agr;.
When it is assumed that the sampling timing is a Nyquist point, that is, when the amplitude component is assumed as
1
, the output (d2(nT)) of the averaging filter
101
is expressed by the following equation (6).
d
2(
nT
)=exp(
jMD&Dgr;&ohgr;T
) (6)
The coordinate converter
24
converts the output (d2(nT)) of the averaging filter
101
from a Cartesian coordinate into a polar coordinate, and calculates the phase component (MD&Dgr;&ohgr;T). Last, the divider
25
divides the phase component (MD&Dgr;&ohgr;T) that is the output of the coordinate converter
24
by MD, thereby to calculate the angular frequency error (&Dgr;&ohgr;T) over one symbol, and outputs a calculated result.
According to the above-described conventional frequency error estimating apparatus, however, in order to estimate a frequency error in high precision, it is necessary to set the coefficient &agr; of the multiplier in the averaging filter to a value as small as possible for increasing the averaging effect. On the other hand, when the received signal receives a large Doppler shift and the Doppler frequency further varies with time like in the mobile communication satellite, it is necessary that the frequency error estimating apparatus follows this variation and estimates the frequency error. In other words, in order to increase this level of following the time variation in the Doppler frequency, it is necessary to set the coefficient &agr; of the multiplier in the averaging filter to a value as large as possible.
As the coefficient &agr; of the multiplier has been fixed in the conventional frequency error estimating apparatus, there has been a problem that it is difficult to satisfy both increasing the precision in estimating the frequency error and increasing the level of following the time variation in the Doppler fre
Bayard Emmanuel
Birch & Stewart Kolasch & Birch, LLP
Mitsubishi Denki & Kabushiki Kaisha
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