Miscellaneous active electrical nonlinear devices – circuits – and – Signal converting – shaping – or generating – Current driver
Reexamination Certificate
2000-09-14
2003-05-27
Callahan, Timothy P. (Department: 2816)
Miscellaneous active electrical nonlinear devices, circuits, and
Signal converting, shaping, or generating
Current driver
C327S112000
Reexamination Certificate
active
06570413
ABSTRACT:
BACKGROUND OF THE INVENTION AND RELATED ART STATEMENT
The present invention relates to a drive circuit for a voltage-controlled switching device, such as insulated-gate bipolar transistor (IGBT), which is used in a power-conversion device, particularly in an inverter for motor speed control, that is, a circuit for accepting external ON/OFF signals as inputs to generate and apply signals for ON/OFF drive directly applied to a control terminal of the voltage-controlled semiconductor switching device. In particular, the invention relates to a drive circuit having a function of minimizing a turn-on time, power dissipation and noise generation in a driven voltage-controlled semiconductor switching element.
In the figures showing below, the same reference signs denote identical or equivalent parts.
FIG. 4
shows an example of a structure of the conventional simplest drive circuit for driving a power switching device, in this case IGBT
1
. In this embodiment, means for charging and turning on the IGBT
1
is a P-channel MOSFET
2
, and means for turning off the IGBT
1
is an N-channel MOSFET
4
.
If an ON signal is input to an ON/OFF-signal input terminal
8
, a predriver
7
first provides an output
7
b
with a low potential (hereinafter also referred to as “L”) to turn off the N-channel MOSFET
4
, and then provides an output
7
a
with “L” to turn on the P-channel MOSFET
2
. When the MOSFET
2
is turned on, it charges the gate capacitance of the IGBT
1
to turn the IGBT
1
ON. This difference in time between the changes in the outputs
7
b
and
7
a
is provided in order to prevent a short-circuit current from flowing when both MOSFETs
2
and
4
are simultaneously turned on due to MOSFET gate delay.
On the other hand, if an OFF signal is input to the ON/OFF-signal input terminal
8
, the predriver
7
first provides the output
7
a
with a high potential (hereinafter also referred to as “H”) to turn off the P-channel MOSFET
2
, and then provides the output
7
b
with “H” to turn on the N-channel MOSFET
4
. Thus, the IGBT
1
gate capacitance is discharged and the IGBT
1
is turned OFF.
In driving the power switching devices, such as IGBTs or MOSFETs, it is important to reduce overall losses including steady-state losses caused by an ON voltage, and switching losses, i.e. turn-on and turn-off losses. If, however, the switching speed is increased to reduce the switching losses, rapid dV/dt or di/dt may cause noise. This noise is normally large when the power switching device is turned on.
In recent years, a great effort has been made in developing drive methods that solve this problem in order to reduce both switching losses and noise. Such drive methods are disclosed in Japanese Patent Application Laid Open (KOKAI) Nos. 61-237513 and 7-240676, wherein a capacitor is connected between the gate and drain of a field-effect transistor (FET) to reduce the ON/OFF speed.
FIG. 5
shows an inverter circuit that uses a pulse-width modulation (PWM) for a motor speed control. The circuit has been simplified to facilitate explanation of the operation of a semiconductor switching device.
FIG. 6
shows the turn-on waveforms of the IGBT
1
in this circuit. An operation is described with reference to
FIGS. 5 and 6
.
First, when the IGBT
1
is turned on, a current flows through a load inductance L. The current increases in terms of di/dt (=Ed/L) determined by the inductance L and a power-supply voltage Ed (specifically, Ed−(ON voltage of the IGBT
1
)=voltage applied to the inductance L).
Next, the IGBT
1
is turned off when a fixed current is reached, and the current flow is commuted through a free-wheeling diode
31
(transition from I
0
to I
1
in FIG.
5
).
Next, when the IGBT
1
is turned on again, all currents flowing through the free-wheeling diode
31
shift to the IGBT
1
at the time that the current through the IGBT
1
has increased to the value of the current through the inductance L (I
1
in FIG.
5
: in fact, this current is slightly less provided by an ON voltage at the free-wheeling diode
31
or the like, but this attenuation is weak since the inductance L is sufficiently large).
FIG. 6
shows the waveforms of a gate voltage V
GE
, a collector current Ic, and a collector-emitter voltage V
CE
observed when the IGBT
1
is turned on under the conditions noted above.
The IGBT
1
is assumed to be driven by the drive circuit in
FIG. 4
, in which the P-channel MOSFET
2
is turned on at time t
1
to start to provide a source current to the gate of the IGBT
1
. When the IGBT gate voltage V
GE
reaches the gate threshold value at time t
2
, collector current Ic starts flowing and increases with the increase of the gate voltage.
When the collector current Ic comes to equal to the forward current through the free-wheeling diode
31
at time t
3
, current no longer flows from the inductance L to the IGBT
1
, because the inductance L is sufficiently large to thereby suppress rapid increase in current flowing through the inductance L.
However, even when no current is flowing through the free-wheeling diode
31
, excess carriers that have been generated by the conductivity modulation while current is flowing remain. If the gate voltage of the IGBT
1
is sufficient to allow a higher collector current Ic to flow, a current (hereinafter referred to as a “reverse recovery current”) I
2
flows transitionally in a direction opposite to a direction where the current was previously flowing through the free-wheeling diode
31
.
This reverse recovery current I
2
flows through the IGBT
1
as shown in
FIG. 5
such that a current flowing through the IGBT
1
is expressed by I
1
(=I
0
)+I
2
. The gate voltage V
GE
of the IGBT
1
attempts to rise continuously due to a charging current from the P-channel MOSFET
2
, but falls steadily along with the collector-emitter voltage V
CE
of the IGBT
1
. This is due to the fact that this voltage drop causes current to flow through a collector-gate capacitance of the IGBT
1
(what is known as the “Miller effect”).
The reason that the collector-emitter voltage V
CE
of the IGBT
1
starts to decline (t
4
) after the reverse recovery current I
2
starts to flow through the free-wheeling diode
31
(t
3
) is now explained. In an area in which the free-wheeling diode
31
can supply a current appropriate for the gate voltage of the IGBT
1
based on a diffusion current associated with a change in an internal carrier distribution, a depletion layer need not extend and the voltage at the free-wheeling diode
31
does not rise, thereby preventing a decrease in the collector-emitter voltage V
CE
of the IGBT
1
.
In an area in which a current can not be supplied unless the reverse recovery current continues to flow through the free-wheeling diode
31
to extend the depletion layer, the reverse recovery current flowing through the free-wheeling diode
31
increases and the collector-emitter voltage V
CE
of the IGBT
1
starts to decrease. Thus, due to the Miller effect, the gate voltage V
GE
of the IGBT
1
falls, and the collector current Ic flowing through the IGBT
1
takes a peak value near time t
4
and subsequently falls to the value of the current Ii previously flowing through the free-wheeling diode
31
. Subsequently, between time t
5
and t
6
, the gate voltage V
GE
remains at or near the value at which the IGBT
1
can maintain the current previously flowing through the free-wheeling diode
31
.
On the other hand, the collector-emitter voltage V
CE
decreases consistently with dV/dt, as shown in FIG.
6
. This is due to the fact that, as the V
CE
diminishes, the current flowing through the collector-gate capacitance of IGBT
1
is balanced by the charging current from the P-channel MOSFET
2
so that they are equal. This also corresponds to a decrease in the collector-emitter voltage V
CE
, which narrows the depletion layer in IGBT
1
and so increases the collector-gate capacitance of the IGBT
1
.
In
FIG. 6
, after time t
5
, the collector current Ic remains constant after the gate voltage V
GE
Kawakami Hiroyuki
Kumagai Naoki
Callahan Timothy P.
Fuji Electric & Co., Ltd.
Kanesaka & Takeuchi
Nguyen Linh
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