Pulse or digital communications – Spread spectrum – Direct sequence
Reexamination Certificate
1998-10-05
2002-02-19
Ghayour, Mohammad H. (Department: 2634)
Pulse or digital communications
Spread spectrum
Direct sequence
C375S143000, C375S142000, C375S150000, C375S152000
Reexamination Certificate
active
06349109
ABSTRACT:
TECHNICAL FIELD
The present invention relates to a direct sequence spread spectrum differential receiver with mixed means for forming an interference signal corresponding to the multiple access noise.
PRIOR ART
The direct sequence spread spectrum modulation technique has been used for many years, particularly in radiocommunications with satellites and in the military sector.
In a digital data emitter using a conventional modulation technique, the data to be emitted modulate a radio-frequency carrier. The modulation used can be a phase, frequency, amplitude or mixed modulation. In order to simplify the description, reference will only be made to phase modulations, which are now the most frequently used.
The digital data to be transmitted consist of binary elements or bits, which have a period T
b
, i.e. a new bit must be transmitted every T
b
. With said bits it is possible to form bit groups, also known as symbols, whose period is T
s
and is a multiple of T
b
. These symbols will modulate the radio-frequency carrier, e.g. in phase.
This technique can be illustrated by two phase modulation examples:
a) The modulation known as binary phase shift keying or BPSK, which consists of allocating a phase state, e.g. 0, to the 0 bits, and a phase state &pgr; to the 1 bits. In this case the symbol is the actual bit (T
s
=T
b
) and the radio-frequency carrier phase state is imposed on every bit.
b) Modulation known as quaternary phase shift keying or QPSK, which consists of using symbols formed by two successive bits, so that said symbols can assume four states (00, 01, 10, 11). A state of the phase of the carrier is allocated to each of these states, in this case T
s
=
2
T
b
and the radio-frequency carrier phase state is imposed on every other bit.
On the reception side, it is necessary to demodulate the signal received. A distinction can be made between two major demodulation families, namely coherent demodulation and non-coherent demodulation. The coherent demodulation technique consists of implementing, in the receiver, a subassembly, whose function is to estimate the mean phase of the carrier, so as to reconstitute a phase reference, which is then mixed with the signal received in order to demodulate the data.
The non-coherent demodulation technique is based on the observation, according to which it is sufficient for the phase reference of the symbol to be compared with the phase of the preceding symbol. In this case, instead of estimating the phase of the symbols, the receiver estimates the phase difference between two successive symbols. This is a differential phase shift keying or DPSK or a differential quadrature phase shift keying or DQPSK.
The attached
FIGS. 1
to
3
diagrammatically show the structure and operation of a spread spectrum emitter and receiver operating in DPSK. This corresponds to FR-A-2 712 129.
FIG. 1
shows the block diagram of an emitter. Said emitter has an input Ee, which receives the data b
k
to be emitted and comprises a differential coder
10
, constituted by a logic circuit
12
and a delay circuit
14
. The emitter also comprises a pseudorandom sequence. generator
30
, a multiplier
32
, a local oscillator
16
and a modulator
18
connected to an output Se, which supplies the DFSK signal.
The logic circuit
12
receives the binary data b
k
and delivers the binary data d
k
. The logic circuit
12
also receives the data delayed by one order or rank, i.e. d
k−1
. The logic operation performed in the circuit
12
is the exclusive-OR on the data b
k
and on the delayed compliment of d
k
(i.e. on {overscore (d
k−1
+L )}):
d
k
=b
k
⊕{overscore (d
k−1
+L )}
The pseudorandom sequence used on emission for modulating the data must have an autocorrelation function with a marked peak (of value N) for a zero delay and the smallest possible secondary lobes. This can be obtained by using maximum length sequences, also called m-sequences, or so-called GOLD or KASAMI sequences in exemplified manner. This pseudorandom sequence designated {c
e
}, has a bit rate N times higher than the rate of the binary data to be transmitted. The duration T
c
of a bit of said pseudorandom sequence and which is also known as a chip is consequently equal to T
b
/N.
The chip rate of the pseudorandom sequence can be several million, or several tens of millions per second.
The attached
FIG. 2
is the block diagram of a corresponding receiver of the differential demodulator type. This receiver has an input Er and comprises a matched filter
20
, whose pulse response is the time reverse of the pseudorandom sequence used in the emitter, a delay circuit
22
with a duration T
b
, a multiplier
24
, an integrator
26
on a period T
b
and a logic decision circuit
28
. The receiver has an output Sr, which restores the data.
If x(t) is used for designating the signal applied to the input Er, the multiplier
24
receives the filtered signal x
F
(t) and the delayed-filtered signal x
F
(t−T
b
) The product is integrated on a period equal to or smaller than T
b
in the integrator
26
, which supplies a signal, whose polarity makes it possible to determine the value of the transmitted bit.
The input filter
20
used in the receiver has a base band equivalent pulse response H(t) and said response must be the time-reverse, conjugate complex of the pseudorandom sequence c(t) used on emission:
H
(
t
)=
c
*(
T
b
−t
)
The signal supplied by such a filter is consequently:
x
F
(
t
)=
x
(
t
)*
H
F
(
t
)
where the symbol * designates the convolution operation, i.e.
x
F
(
t
)
=
∫
0
T
b
x
(
s
)
·
c
*
(
s
-
t
)
ⅆ
s
.
Thus, the matched filter
20
performs the correlation between the signal applied at its input and the pseudorandom spread sequence.
In a gaussian additive noise channel, the signal x(Ft) will consequently be in the form of a pulse signal, the pulse repetition frequency being 1/T
b
. The envelope of this signal is the autocorrelation function of the signal c(t). The information is carried by the phase difference between two successive correlation peaks. Thus, the multiplier output is formed by a succession of positive or negative peaks, as a function of the value of the transmitted bit.
In the case of a radiotransmission in the presence of multiple paths, the output of the matched filter is formed by a succession of correlation peaks, each peak corresponding to a propagation path.
The different signals of the reception chain are represented in FIG.
3
.
Line (a) represents the filtered signal x
F
(t), line (b) the correlation signal x
F
t)*x
F
(t−T
b
) and line (c) the signal at the integrator output.
The direct sequence spread spectrum modulation technique has been extensively described in the specialist literature and reference can e.g. be made to the following works:
“CDMA Principles of Spread Spectrum Communication”, by Andrew J. VITERBI, Addison-Wesley Wireless Communications Series,
“Spread Spectrum Communications”, by Marvin K. SIMON et al., vol. I, 1983, Computer Science Press,
“Spread Spectrum Systems”, by R. C. DIXON, John WILEY and Sons.
This technique is also described in certain articles;
“Direct-sequence Spread Spectrum with DPSK Modulation and Diversity for Indoor Wireless Communications”, published by Mohsen KAVEHRAD and Bhaskar RAMAMURTHI in the journal “IEEE Transactions on Communications”, vol. COM 35, No. 2, February 1987,
“Practical Surface Acoustic Wave Devices”, by Melvin G. HOLLAND, in the journal Proceedings of the IEEE, vol. 62, No. 5, May 1974, pp 582-611.
The direct sequence spread spectrum technique has numerous advantages, such as:
Discretion; this discretion is linked with the spread of the transmitted information over a wide frequency band, leading to a low spectral density of the emitted power.
Multiple access: several direct sequence spread spectrum links can share the same frequency band using orthogonal spread pseudorandom sequences (sequences having an intercorrelation function having very low residual noise for all shift
Daniele Norbert
Lattard Didier
Lequepeys Jean-Rene
Varreau Didier
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