Digital AM demodulator, particularly for demodulating TV...

Details Digital AM demodulator, particularly for demodulating TV... Digital AM demodulator, particularly for demodulating TV...

2000-05-16

2004-06-08

Chin, Stephen (Department: 2734)

Pulse or digital communications

Receivers

Amplitude modulation

C455S255000

Reexamination Certificate

active

06748028

ABSTRACT:

FIELD OF THE INVENTION
The present invention relates to a digital AM demodulator, and, more particularly, to a high-frequency digital AM demodulator adapted for demodulating TV signals in PAL-NTSC-SECAM formats, which can be integrated in a digital video processing unit. The digital AM demodulator starts with a medium-frequency sampled signal, such as a signal originating from a tuner.
BACKGROUND OF THE INVENTION
To demodulate a TV signal starting from a medium-frequency sampled signal originating from a tuner, for example, an analog demodulator is positioned downstream from the tuner. The output is a signal that is demodulated in an audio demodulator to obtain an audio signal, and a video signal that is split into a luminance signal and a chrominance signal.
We now consider a known AM demodulator, a block diagram of which is shown in FIG.
1
. The demodulator is formed by an input filter
1
and by a low-pass filter
2
arranged in a cascade configuration. The input filter
1
is fed with an input signal and the output signal therefrom is fed to a multiplier
3
, which in turn receives a local carrier signal.
Assuming, therefore, that a local carrier designated by the term “carrier” is available, whose value is
carrier=2 cos(&ohgr;
a
t
+&psgr;),
and according to trigonometric addition and subtraction formulas
cos
⁡
(
α
)
+
cos
⁡
(
β
)
=
cos
⁢
(
α
+
β
)
2
·
cos
⁢
(
α
+
β
)
2
the carrier (carrier) and the received signal x(t) produce the demodulation of A(t) according to the following relations:
input
x=A
(
t
)·cos(&ohgr;
i
t
+&phgr;)
carrier=2·cos(&ohgr;
a
t
+&psgr;)
y
t
=input·carrier
y
t
=2
·A
(
t
)·cos(&ohgr;
i
t
+&phgr;)·cos(&ohgr;
a
t
+&psgr;)=
A
(
t
)·cos[(&ohgr;
i
+&ohgr;
a
)
t
+(&phgr;+&psgr;)]+cos[(&ohgr;
i
−&ohgr;
a
)
t
+(&phgr;−&psgr;)]
After multiplication, one obtains a frequency composed of two components, as shown in FIG.
3
. One is modulated around the frequency (&ohgr;
i
−&ohgr;
a
), the other one is modulated around the frequency (&ohgr;
i
+&ohgr;
a
). If the higher-frequency component is removed with a low-pass filter and the condition &ohgr;
i
=&ohgr;
a
is set, one obtains:
y=A
(
t
)·cos[(&ohgr;
i
−&ohgr;
a
)
t
+(&phgr;−&psgr;)=
A
(
t
)·cos(&phgr;−&psgr;)
t=A
(
t
)·
k
(
t
)
w
i
=&ohgr;
a
|k|≦
1
Since the coefficient k has a modulus of 1 or less, and to also provide the demodulated channel with the maximum energy level (SNR=max, where SNR is the signal
oise ratio), it is necessary to set:
&ohgr;
a
=&ohgr;
i
=>y=A
(
t
)·cos(&phgr;−&psgr;)
&phgr;=&psgr;=>
y=A
(
t
)
Using the same reasoning and with reference to
FIGS. 3 and 4
, it can be easily demonstrated that it is possible to perform baseband demodulation of the transposed input channel or of its exact symmetrical counterpart by acting on the position of the frequency of the local carrier. If f
c
=f
i
the baseband spectrum is the input spectrum, and if f
c
=f
h
; then a symmetrical spectrum is obtained.
The above is a direct demodulation. We now describe an AM demodulation achieved in two steps, i.e., with the aid of two carriers, designated by carrier
1
and carrier
2
. With reference to the diagram of
FIG. 2
, the reference numeral
1
designates, as in the preceding case, an input filter, whereas the reference numeral
2
designates the low-pass filter and
3
is the multiplier to which the first carrier, carrier
1
, is fed. The diagram of
FIG. 2
provides for a second multiplier
4
to which the second carrier, carrier
2
, is fed and an additional low-pass filter
5
is arranged downstream of the multiplier
4
.
In this case, one has the following relations:
input=
A
(
t
)·cos(&ohgr;
i
t
+&phgr;)
carrier
1
=2·cos(&ohgr;
a
t
+&psgr;)
carrier
2
=2·cos(&ohgr;
b
t
+&phgr;)
(&ohgr;
a
+&ohgr;
b
)=&ohgr;
i
y
a
=input·carrier
1
 
y
a
=2
·A
(
t
)·cos(&ohgr;
i
t
+&phgr;)·cos(&ohgr;
a
t+y
)=
A
(
t
){cos[(&ohgr;
i
+&ohgr;
a
)
t
+(&phgr;+&psgr;)]+cos[(&ohgr;
i
−&ohgr;
a
)
t
+(&phgr;−&psgr;)]}
After eliminating the high-frequency component (&ohgr;
i
+&ohgr;
a
) one obtains:
y
b
=A
(
t
)·cos[(&ohgr;
i
−&ohgr;
a
)
t
+(&phgr;+&psgr;)]
After the first multiplication in the first multiplier 3, the spectrum A(t) is modulated around the intermediate frequency &ohgr;
int
=(&ohgr;
i
−&ohgr;
a
), with the initial phase f
int
=(&phgr;−&psgr;). The spectrum has been shifted to a lower frequency. With the second stage, shown in
FIG. 2
, one obtains:
y
c
=y
b
·carrier
2
y
c
=2
·A
(
t
)·cos(
w
int
t+&phgr;
int
)·cos(&ohgr;
b
t
+&phgr;)
y
c
=A
(
t
){cos[(
w
int
+&ohgr;
b
)
t
+(&phgr;
int
+&phgr;)]+cos[(&ohgr;
int
−&ohgr;
b
)
t
+(&phgr;
int
−&phgr;)]}
and after the low-pass filter
5
y=A
(
t
)·cos[(&ohgr;
int
−&ohgr;
b
)
t
+(&phgr;
int
−&phgr;)]
The conditions under which the input signal A(t) is correctly baseband-demodulated are as follows:
&ohgr;
i
=(&ohgr;
a
+&ohgr;
b
)  condition 1
&phgr;=(&psgr;+&phgr;)  condition 2
It can be noted that both the frequency &ohgr;
i
and the phase &phgr; can be distributed at will between the two local carriers, carrier, and carrier
2
. Accordingly, the following problems arise in the design of a demodulator, particularly of the digital type.
Since at the output of a tuner the phase &ohgr; is not known in advance and the medium-frequency spectrum has a frequency shift which can vary by a few hundred kHz with respect to its nominal value (for a video tuner the band is 33-38.9 MHZ+/−100 kHz), the need for a PLL is evident for analog recovery signals to extract &ohgr;
i
in phase with the received signal f. For medium-high frequencies (f>20 MHz), digital PLLs are typically more burdensome than analog equivalents for an equal performance, and are therefore rarely used.
Another drawback is noted with discrete-time systems where t=nTs, with T equal to the sampling period. In this case, the frequency spectrum is periodic, with a period 2p, and this can cause an unwanted aliasing effect during demodulation, as shown in
FIG. 5
b.
In
FIG. 5
a,
the carrier lies at the vertex B of the input spectrum and demodulation occurs correctly. In
FIG. 5
b,
the carrier lies in the upper part of the spectrum, at the vertex A. In this case, the first two repeats of the sampled input spectrum are demodulated, distorting the band signal. This effect is not observed in the case of analog demodulation (see
FIGS. 3 and 4
) and forces a two-step demodulation, as shown in
FIG. 5
c.
In this case, the first demodulation is meant to shift the input spectrum to a lower frequency and to shift the second demodulation to a higher frequency. With the second demodulation, the channel is brought to baseband correctly without aliasing errors, but this nearly doubles the complexity of the architecture, which uses at least two multipliers and three filters, as shown in FIG.
2
.
SUMMARY OF THE INVENTION
An object of the present invention is to provide a digital AM demodulator in which recovery of the frequency and phase of the carrier is performed without resorting to a PLL.
A further object of the present invention is to provide a digital AM demodulator which can be easily integrated in digital video decoders.
Another object of the present invention is to provide a digital AM demodulator in which the complexity of the circuit and the occupied area are reduced with respect to known approaches.
Yet another object of the present invention is to provide a digital AM demodulator whi

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