Pulse or digital communications – Receivers – Angle modulation
Reexamination Certificate
1999-02-01
2002-09-17
Chin, Stephen (Department: 2634)
Pulse or digital communications
Receivers
Angle modulation
C375S262000, C375S285000, C375S326000, C375S341000, C375S346000
Reexamination Certificate
active
06452983
ABSTRACT:
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to digital modulation/demodulation used for satellite communication, mobile communication, mobile satellite communication, in particular, to a demodulator periodically inserting a known signal into transmitted signals, estimating channel distortion from the known signal, eliminating the channel distortion using the estimated value, and performing coherent detection of the received signal.
2. Description of the Related Art
Recently, in the field of satellite communication, mobile communication, and mobile satellite communication, an investigation of digital modulation/demodulation has been actively performed. Especially, in environments of the mobile communication, a signal is received with fading. Various demodulation systems are examined, which stably operates under such fading environment. Among these systems, a system is remarkably noticed as a system capable to perform absolute coherent detection under fading environment, in which a known signal is periodically inserted into transmitted signals for calibration of channel distortion such as fading, and estimation and compensation of fading distortion is performed based on the known signal.
FIGS. 22 and 23
show conventional demodulation circuits disclosed in, for example, “Rayleigh Fading Compensation Method for 16QAM MODEM in Digital Land Mobile Radio Systems” (Sampei, Theses of The Institute of Electronics, Information and Communication Engineers (B-II), J72-II, No. 1, pp7-15 (1989-1)). In
FIG. 22
, a reference numeral
40
shows a quasi-coherent detecting unit, a reference numeral
50
p
shows a carrier estimator and
60
shows a data discriminator. In
FIG. 23
, a reference numeral
51
shows a fading distortion estimator and
52
shows a fading distortion compensator.
In the following, an operation of the above demodulation circuit will be explained.
Here, it is assumed a signal is modulated by QPSK (Quadriphase Phase Shift Keying). A transmitted signal S
T
(t), modulated by QPSK, is given by the following expression (1).
In expression (1), Re[•] shows a real part of [•] and f
c
shows a carrier frequency. z
T
(t) shows a transmitted baseband signal and is given by expression (2).
S
T
(
t
)=
Re[z
T
(
t
)exp(
j
2
&pgr;f
c
t
)] (1)
z
T
(
t
)=
z
p
(
t
)+
j·z
Q
(
t
) (2)
In case of the above signal transmitted under the fading environment, the faded signal is represented by a narrow-band random complex signal c(t) having fading power spectrum S(f) multiplied to S
T
(t). The received signal S
R
(t) faded by an envelope variation c(t) is given by expression (3).
S
R
(
t
)=
Re[c
(
t
)
z
R
(
t
)exp(
j
2
&pgr;f
c
t
)+
n
(
t
)exp(
j
2
&pgr;f
c
t
)] (3)
In the above expression (3), z
R
(t) denotes the received baseband complex signal limited by the transmitter and receiver filters. n(t) denotes a white Gaussian noise.
In the following, the receiver will be discussed.
The quasi-coherent detection is performed on the received signal by the quasi-coherent detecting unit
40
with a local oscillator having oscillating frequency of f
c
−f
off
. A signal S
RB
(t) output from the quasi-coherent detecting unit
40
is given by the following expression (4).
s
RB
(
t
)=
Re└u
(
t
)exp(
j
2
&pgr;f
off
t
)┘
u
(
t
)=
c
(
t
)
z
R
(
t
)+
n
(
t
) (4)
The carrier estimator
50
p
obtains c
R
(t), the estimation value of c(t) from the baseband signal u(t) output from the quasi-coherent detecting unit
40
. The carrier estimator
50
p
also eliminates the distortion caused by c(t) from the received signal. Here, it is assumed that the frequency offset value f
off
is sufficiently small (f
off
=0). When the known signal is supposed to be a
p
, z
R
(t) becomes z
R
(t)=a
p
. Therefore, the received baseband complex signal u(t) is given by the following expression (5).
Accordingly, c
R
(t), the estimation value of c(t) is obtained by the expression (6).
u
(
t
)
=
c
(
t
)
a
p
+
n
(
t
)
(
5
)
c
R
(
t
)
=
1
a
p
u
(
t
)
=
c
(
t
)
+
1
a
p
n
(
t
)
(
6
)
Based on c
R
(t) estimated for each known signal, the estimation sequence c
RS
(t) is represented by the following expression (7) when an insertion interval of the known signal is T
R
.
c
RS
(
t
)
=
∑
k
=
-
∞
∞
c
R
(
k
)
δ
(
t
-
k
T
R
)
(
7
)
An estimation of an interpolation sequence c
RI
(t) out of c
RS
(t) is, for example, performed by the following way using the interpolation.
In the estimation of fading distortion using the interpolation, the fading distortion of the information signal among the known signal is estimated by the interpolation using the estimation value of the fading distortion obtained by the known signal. Namely, in the fading distortion estimator
51
estimates the fading distortion c
R
(k−1), c
R
(k), c
R
(k+1), obtained at t=(k−1)T
R
, kT
R
, (k+1)T
R
corresponding to the known signal among the signals output from the quasi-coherent detecting unit
40
as shown in FIG.
24
. The fading distortion at t=kT
R
+(m/N
R
) T
R
(N
R
: the number of symbols corresponding to the insertion interval of the known signal) is estimated, for example, using the second order Gaussian interpolation and given by the following expression (8).
c
RI
(
k
+
m
N
R
)
=
Q
-
1
(
m
N
R
)
c
R
(
k
-
1
)
+
Q
0
(
m
N
R
)
c
R
(
k
)
+
Q
1
(
m
N
R
)
c
R
(
k
+
1
)
Q
-
1
(
m
N
R
)
=
1
2
{
(
m
N
R
)
2
-
m
N
R
}
Q
0
(
m
N
R
)
=
1
-
(
m
N
R
)
2
Q
1
(
m
N
R
)
=
1
2
{
(
m
N
R
)
2
+
m
N
R
}
}
(
8
)
The fading compensator
52
eliminates the fading distortion from the received signal using the interpolation sequence c
RI
(k+m/N
R
) output from the fading distortion estimator
51
. Namely, a transmission function h(k+m/N
R
) for compensating the fading distortion is represented by the following expression (9).
From the expression (9), the fading distortion of the signal output from the quasi-coherent detecting unit
40
is compensated as shown in the expression (10).
h
(
k
+
m
N
R
)
=
1
c
RI
(
k
+
m
N
R
)
(
9
)
z
RE
(
k
+
m
N
R
)
=
h
(
k
+
m
N
R
)
·
u
(
k
+
m
N
R
)
=
u
(
k
+
m
N
R
)
c
RI
(
k
+
m
N
R
)
(
10
)
As described above, the received signal z
RE
(k+m/N
R
), of which fading distortion was compensated, can be obtained.
According to the conventional fading distortion estimation and compensation method, the fading distortion of the information signal is estimated by the interpolation using the estimation value of the fading distortion obtained by the known signal. Therefore, in the nonfading channel or Rician fading channel, where thermal noise influences a lot in addition to the fading variation, an error of the estimation value obtained by the known signal becomes large, the fading distortion cannot be compensated properly, which degrades performance of the receiver such as a bit error rate characteristic.
Further, even in the fading channel, thermal noise influences a lot when C/N (carrier power to noise power rate) is low, and the estimation error of the fading distortion estimated by the known signal becomes large. The fading distortion cannot be compensated properly, which degrades the performance of the receiver such as the bit error rate characteristic.
Further, the fading channel and the nonfading channel have opposite characteristics, so that when the performance of the receiver is improved in one of two above channels, the performance of the receiver is degraded in the other channel. It is difficult to embody the receiver having a high performance in wide range from the fading channel to nonfading channel.
SUMMARY OF THE INVENTION
The present invention is provided to solve the abovementioned problems. The invention aims to have a demodulator, where the estima
Asahara Takashi
Miyake Makoto
Chin Stephen
Fan Chieh M.
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