Electric power conversion systems – Current conversion – Including d.c.-a.c.-d.c. converter
Reexamination Certificate
1999-12-23
2001-02-27
Riley, Shawn (Department: 2838)
Electric power conversion systems
Current conversion
Including d.c.-a.c.-d.c. converter
Reexamination Certificate
active
06195273
ABSTRACT:
FIELD OF THE INVENTION
The present invention relates generally to DC:DC power converters, and more specifically to providing a converter topology that improves efficiency, reduces EMI and optionally provides a second output voltage that is half the main output voltage.
BACKGROUND OF THE INVENTION
Circuitry to implement DC:DC converters is known in the art. Such circuits receive an input-side DC voltage that is sampled or chopped and transformer-coupled to an output side. On the output side, the waveform is rectified and filtered to provide a regulated output voltage that may be greater than or less than the input voltage. Feedback from output to input can be used to regulate the sampling duty cycle or frequency to provide an acceptably efficient DC:DC converter in a small form factor.
FIG. 1A
depicts a so-called voltage-fed push-pull DC:DC converter
10
, according to the prior art, has having an input or primary side
20
and an output or secondary side
30
. In input and output sides are essentially demarked by a transformer T
1
having input windings W
1
, W
2
, and output windings W
3
-
1
,W
3
-
2
and W
4
-
1
,W
4
-
2
. Preferably windings W
1
and W
2
are identical and center tapped windings W
3
-
1
,W
3
-
2
,W
4
-
1
,W
4
-
2
are identical.
The input side
20
of the converter is coupled to a source of DC potential Vin that in some applications may be pre-regulated with a pre-regulator
40
whose output potential is controlled within a known tolerance. In other applications, preregulation is omitted and feedback
50
is used to modulate pulse width on drive signals output from a control circuit
60
, to regulate the output voltage(s), shown here as V
01
, V
02
.
In
FIG. 1A
, input voltage, which may be the output potential from pre-regulator
30
, is sampled or chopped using push-pull switching transistors Q
1
, Q
2
and respective transformer T
1
primary windings W
1
, W
2
. Control circuit
50
provides complementary drive signals to the input leads of Q
1
, Q
2
such that when Q
1
is on, Q
2
is off, and vice versa. Although Q
1
and Q
2
are shown as switching an end of primary windings W
1
, W
2
to ground potential, it is understood that ground potential implies a stable potential. Stated differently, if desired a potential other than 0 V DC might instead be switchably coupled to an end of primary windings W
1
and W
2
. This understanding that ground is simply a convenient reference potential shall apply throughout this disclosure.
On the converter output side
30
, center-tapped secondaries W
3
-
1
, W
3
-
2
, and W
4
-
1
, W
4
-
2
of transformer T
1
step-up or step-down the chopped waveforms, which are rectified by diodes D
1
, D
2
and inductor L
1
-capacitor C
1
, and by diodes D
3
, D
4
and inductor L
2
-capacitor C
2
. As described below, in an attempt to reduce voltage stress on the output side rectifier components and to reduce EMI it is customary to insert snubbers, typically a series-coupled resistor-capacitor, across each output winding of T
1
.
Feedback loop
50
can sample the DC output voltages, here shown as Vo1, Vo2, to control the pulse width (or duty cycle) and/or frequency of the Q
1
, Q
2
drive signals generated by control circuit
60
. The secondary windings may output different magnitudes Vo1, Vo2 and the number of windings may be greater or less than two.
As will be described in detail shortly, there are several recognized drawbacks with the configuration of
FIG. 1A
, including difficulty in implementing transformer T
1
, under utilization of the secondary transformer windings, the essentially unbalanced state of the magnetic flux in the transformer, and the need to employ snubbers to help protect against voltage stress and reduce EMI, at the cost of conversion efficiency. A further drawback to the configuration of
FIG. 1A
is the necessity to ensure that Q
1
and Q
2
are never simultaneously in the on-state, a condition that could result in potentially destructive inrush current levels. High current transients during any overlap between transition states of Q
1
, Q
2
gives rise to high magnitude of electromagnetic (EMI) radiation, which can require expensive shielding of the DC:DC converter. Ensuring that Q
1
and Q
2
are not simultaneously on can add to the complexity of control circuit
60
.
In the push-pull configuration of
FIG. 1A
(and indeed
FIG. 1B
as well) careful flux matching is required for transformer T
1
, especially balancing between windings W
1
and W
2
and switches Q
1
, Q
1
. On the output side, each of D
1
and D
2
, and D
3
and D
4
deliver the same amount of energy to their respective loads (not shown). This in turn dictates good symmetry between center-tapped windings W
3
-
1
and W
3
-
2
, and W
4
-
1
and W
4
-
2
. Such symmetry can prevent or at least greatly reduce the presence of harmonic energy at half the switching frequency of Q
1
or Q
2
. Understandably, implementing a well designed voltage-fed pushpull DC:DC converter can be challenging.
Fabricating a perfectly symmetrical transformer T
1
is difficult in practice. But even if the T
1
windings are perfectly balanced, the secondary windings are never fully utilized in the sense that these windings do not conduct current all of the time. For example, if D
1
is conducting, winding W
3
-
1
is used (e.g., conducts current), but during this time diode D
2
is not conducting and winding W
3
-
2
is not used. By the same token, if D
2
is conducting, winding W
3
-
2
is used, but winding W
3
-
1
conducts no current and is not used. (The same statements hold true for windings W
4
-
1
, W
4
-
2
, if they are present).
This under utilization of the secondary transformer windings presents several problems. During the time a secondary transformer winding is not being used, the winding portion coupled to the associated (reverse-biased) diode is essentially floating, e.g., not clamped to a low impedance. As a result voltage spikes can be generated, which give rise to overshoot and undershoot ringing and EMI, which can be unacceptable in many applications. Further, the voltage spikes can overstress the rectifier diode, inductor, and output capacitors, requiring higher voltage breakdown components to be used, thus increasing cost and perhaps size of the circuitry.
Those skilled in the art will appreciate that coupling an R-C snubber across the transformer secondary windings presents a lower AC-impedance that can dampen the magnitude of otherwise dangerous voltage spikes, thus reducing EMI. Unfortunately, the snubbers dissipates inductance leakage energy, and can reduce DC:DC conversion by up to 5% or so. However where EMI and over-voltage stress present critical constraints, prior art configuration
1
A will require snubbers. In some instances, it may even be necessary to reduce switching speed in prior art converters in an attempt to control voltage stress and EMI generation. If the snubbers were eliminated, conversion efficiency would increase by several percent but the magnitude of the overshoot/undershoot could be 100% of the voltage level being switched. Thus if Vo1 were 50 VDC, capacitor C
1
would have to be rated at at least 100 VDC breakdown. If it were somehow possible to more fully or more efficiently utilize transformer
40
, it might be possible to use a smaller transformer, e.g., a lighter and less expensive transformer.
FIG. 1B
depicts a so-called current-input topology for a push-pull DC:DC converter system
80
. Components similarly numbered as in
FIG. 1A
may generally be the same as those described with respect to converter system
10
. In this configuration, Vin (which may be pre-regulated) is switched via Qo (under command of control circuit
90
) to lowpass filter Lo-Co. The filtered Vin is then switchingly coupled to T
1
primary windings W
1
, W
2
by switches Q
1
, Q
2
under command of control circuit
90
. Since input switch Qo is sometimes open, a diode Do is included to ensure a current path for Lo when Qo is open. In the configuration of
FIG. 1B
, drive signals to
01
and Q
2
are never on simultaneously.
A current-
Flehr Hohbach Test Albritton & Herbert LLP
Riley Shawn
Switch Power, Inc.
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