Control circuit for LED and corresponding operating method

Details Control circuit for LED and corresponding operating method Control circuit for LED and corresponding operating method

2001-02-12

2002-06-04

Wong, Don (Department: 2821)

Electric lamp and discharge devices: systems

Current and/or voltage regulation

C315S18500S, C315S224000

Reexamination Certificate

active

06400101

ABSTRACT:

The invention is based on a drive circuit for LEDs and an associated operating method as claimed in the preamble of claim 1. This relates in particular to reducing the drive power losses in light-emitting diodes (LEDs) by means of a pulsed LED drive circuit.
As a rule, series resistors are used for current limiting when driving light-emitting diodes (LEDs), see, for example, U.S. Pat. No. 5,907,569. A typical voltage drop across light-emitting diodes (U
F
) is a few volts (for example, for Power TOPLED U
F
=2.1 V). The known resistor R
v
in series with the LED (see
FIG. 1
) produces a particularly high power loss, particularly if the battery voltage U
Batt
is subject to major voltage fluctuations (as is normal in motor vehicles). The voltage drop across the LEDs still remains constant even when such voltage fluctuations occur, that is to say the residual voltage across the series resistor R
v
falls. R
v
is thus alternately loaded to a greater or lesser extent. In practice, a number of LEDs are generally connected in series (in a cluster) in order to achieve better drive efficiency (FIG.
2
). Depending on the vehicle power supply system (12 V or 42 V), a large number of LEDs can accordingly be combined to form a cluster. With a 12 V vehicle power supply system, there is a lower limit on the battery voltage U
Batt
down to which legally specified safety devices (for example the hazard warning system) must be functional. This is 9 volts. This means that, in this case, up to four Power TOPLEDs can be combined to form a cluster (4×2.1 V=8.4 V).
The power loss in the series resistor is converted into heat, which leads to additional heating—in addition to the natural heating from the LEDs in the cluster.
The technical problem is to eliminate the additional heating (drive power loss from the series resistors). There are a number of reasons for this. Firstly, enormous losses occur in the series resistor; in relatively large LED arrays, this can lead to a power loss of several watts. Secondly, this heating from series resistors itself restricts the operating range of the LEDs. If the ambient temperature T
A
is high, the maximum forward current I
F
=f (T
A
) must be reduced in order to protect the LEDs against destruction. This means that the maximum forward current I
F
must not be kept constant over the entire ambient temperature range from 0 to 100° C. In addition, when LEDs with series resistors are being operated, another problem is the fluctuating supply voltage, as is frequently the case in motor vehicles (fluctuation from 8 to 16 V with a 12 V power supply system; fluctuation from 30 to 60 V with the future 42 V vehicle power supply system). Fluctuating supply voltages lead to fluctuating forward currents I
F
, which then result in different light intensities and, associated with this, fluctuations in the brightness of the LEDs.
In the past, series resistors have always been used to limit the forward current through the LEDs. In most cases, the same board has been used for all the series resistors and, if possible, this has been mounted at a suitable distance from the LEDs. This distance was chosen so that the heating from the series resistors R
v
did not influence the temperature of the LEDs.
A further problem is the choice of the maximum forward current I
F
of LEDs. When operating LEDs with series resistors R
v
, the maximum permissible forward current I
F
cannot be chosen, since the forward current must be reduced if the ambient temperature T
A
is higher. A forward current I
F
is therefore chosen which is less than the maximum permissible current (FIG.
3
). This admittedly increases the temperature range for operation of the LEDs, but does not utilize the forward current I
F
optimally. The example in
FIG. 3
(Power TOPLED, Type LA E675 from Siemens) shows the forward current I
F
as a function of the ambient temperature T
A
. The maximum forward current I
F
may in this case be 70 mA up to an ambient temperature of 70° C. Above an ambient temperature of 70° C., the forward current I
F
must then be reduced linearly, until it is only 25 mA at the maximum permissible ambient temperature of 100° C. A variable series resistor R
v
would have to be used for optimum utilization of this method of operation of LEDs.
A further problem is voltage fluctuations. Until now, there have been no drive circuits for LEDs in practical use in order to prevent voltage fluctuations, and thus forward-current fluctuations (brightness fluctuations). They therefore have had to be tolerated by necessity.
The object of the present invention is to provide a drive circuit for an LED as claimed in the preamble of claim 1, which produces as little emitted heat and power loss as possible.
This object is achieved by the distinguishing features of claim 1. Particularly advantageous refinements can be found in the dependent claims.
A pulsed LED drive is used in order to eliminate the series resistor R
v
and thus the high drive power loss.
FIG. 4
a
shows the principle of pulsed current regulation for LEDs. A semiconductor switch, for example a current-limiting power switch or, preferably, a transistor T (in particular of the pnp type, although the npn type is also suitable if a charging pump is also used for the drive), is connected by its emitter to the supply voltage (U
Batt
) (in particular the battery voltage in a motor vehicle) . When the transistor T is switched on, a current i
LED
flows through the LED cluster (which, by way of example, in this case comprises four LEDs), to be precise until the transistor T is switched off again by a comparator. The output of the comparator is connected to the base of the transistor. The one (positive) input of the comparator is connected to a regulation voltage, and the second (negative) input of the comparator is connected to a frequency generator (preferably a triangle waveform generator with a pulse duration T
p
and, accordingly, a frequency 1/T
p
, since this has particularly good electromagnetic compatibility, although other pulse waveforms such as a sawtooth are also possible). The transistor T is switched on if the instantaneous amplitude of the triangle waveform voltage U
D
at the comparator is greater than the regulation voltage U
Reg
. The current which flows is i
LED
. When the instantaneous amplitude of the triangle waveform voltage falls below the constant value of the regulation voltage U
Reg
on the comparator, the transistor T is switched off again. This cycle is repeated regularly at the frequency f at which the triangle waveform generator operates.
The current flowing via the LEDs is pulsed in this way (
FIG. 4
b
). The square-wave pulses have a pulse width which corresponds to a fraction of T
p
. The interval between the rising edges of two pulses corresponds to T
p
.
The LEDs are connected in series with a means for measuring the current (in particular a measurement resistor R
Shunt
between the LEDs and ground (case 1) or else between the semiconductor switch (transistor T) and the terminal of the supply voltage U
Batt
(case 2)). The pulsed current i
LED
is tapped off on the measurement resistor R
Shunt
. The mean value of the current {overscore (i)}
LED
is then formed via an auxiliary means. The auxiliary means is, for example, an integration means (in case 1), preferably an RC low-pass filter, or a differential amplifier (in case 2). This mean value is used as the actual value for current regulation, and is provided as an input value to a regulator (for example a PI or PID regulator). A nominal value, in the form of a reference voltage (U
Ref
) for current regulation is likewise provided as a second input value to the regulator. The regulation voltage U
Reg
at the output of the regulator is set by the regulator such that the actual value always corresponds as well as possible to the nominal value (in terms of voltage). If the supply voltage U
Batt
varies due to fluctuations, the on-time of the transistor T and the length of the square-wave pulse (
FIG. 4
b
) are also adapted as appropriate. This

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