Oscillators – Automatic frequency stabilization using a phase or frequency... – Particular error voltage control
Reexamination Certificate
2002-06-14
2004-04-20
Nguyen, Minh (Department: 2816)
Oscillators
Automatic frequency stabilization using a phase or frequency...
Particular error voltage control
C331S016000, C327S105000, C332S128000
Reexamination Certificate
active
06724265
ABSTRACT:
FIELD OF THE INVENTION
The present invention relates in general to frequency synthesizers, and in particular, to a technique to compensate for tuning gain variations.
BACKGROUND OF THE INVENTION
With reference to
FIG. 1
, an exemplary fractional-N frequency synthesizer
12
is illustrated according to one embodiment of the present invention. The synthesizer
12
includes a fractional-N phase lock loop (PLL)
14
, and generates a desired frequency for the output signal, F
VCO
,
16
, of a voltage controlled oscillator (VCO)
18
. In traditional fashion, the output signal F
VCO
16
is also provided to divider circuitry
20
to divide the output signal F
VCO
by a factor N to produce a divided VCO signal F
V
, which is fed to one of two inputs of a phase detector
22
.
A reference signal, F
REF
, is divided by a factor R in divider circuitry
24
to produce a divided reference signal, F
r
, which is provided to the other input of the phase detector
22
. The N and R factors are selected so that the frequencies of the divided reference signal, F
r
, and the divided VCO signal, F
V
are equal when the desired output signal, F
VCO
,
16
, is at a desired frequency. The phase detector
22
compares the relative phases of the divided reference signal, F
r
, and the divided VCO signal, F
V
, and provides an output relative to the difference in phase to drive a charge pump
26
.
The phase detector
22
is typically an asynchronous digital logic circuit that pulses either pump up (PU) or pump down (PD) signals for the duration of time between rising edges on the reference signal, F
r
, and divided VCO signal, F
V
. The PU and PD signals cause the charge pump
26
to source or sink current, ICP, from a low pass filter, generally referred to as the loop filter
28
. The loop filter
28
is typically a passive or active RC filter, and the one or more pulses of current are integrated and stored on the loop filter's capacitors as charge. The output voltage of the loop filter
28
is a function of this charge, and acts as the tuning control voltage V
CON
of the VCO
18
. The N divider circuitry
20
is typically a programmable integer or fractional divider, which is used to set the output frequency of the VCO
18
. The PLL
14
acts as a feedback control system to drive the phase, and therefore frequency, error of the F
r
and F
v
signals to zero. Since F
v
=F
VCO
/N, where N is the divider modulus, the VCO frequency is set to F
VCO
=N F
r
.
The behavior of the PLL
14
in terms of noise and dynamic response is determined by the loop gain of the system. The loop gain is given by:
G
(
s
)
=
I
CP
K
v
F
(
s
)
sN
,
Eq
.
1
where s is the Laplace frequency variable, I
CP
is the charge pump current in amperes (A), K
V
is the tuning gain in cycles-per-second-per-volt (Hz/V), F(s) is the loop filter transfer function, and N is the VCO divider modulus. A typical filter transfer function contains a gain set by a capacitance, or combination of capacitances, an integration function, and a lead-lag pole/zero combination to set the phase margin of the loop:
F
(
s
)
=
1
sC
(
s
τ
z
+
1
)
(
s
τ
p
+
1
)
.
Eq
.
2
The unity loop gain frequency, also referred to as the loop bandwidth, is given by:
BW
=
I
CP
K
v
K
f
NC
Eq
.
3
where the variable, K
f
, is a factor that depends on the locations of the poles and zeros. Note that the loop bandwidth depends not only on the pole and zero locations, but also on the loop gain constant set by the charge pump current, I
CP
, the loop divider value, the filter capacitance, and the VCO tuning gain.
Modern communication systems, such as the GSM cellular telephone system, impose strict requirements on the locktime and noise performance of the transmitted signal, and on the signals used for mixing in the receiver. For example, the transmit locktime must typically be under 250 &mgr;s to settle the VCO to under 100 Hz error, and the transmitted phase noise must be under—113 dBc/Hz at 400 kHz offset. If the loop bandwidth is too wide, the noise performance may not be met, and if the loop bandwidth is too narrow, the locktime may not be met. In addition, the phase error of the transmitted signal must remain small. For example, the phase error of the transmitted signal must be under five degrees rms in a GSM system.
The use of fractional-N synthesis enables digital modulation for phase or frequency based systems and is attractive due to reduced complexity of the transmit system relative to traditional analog modulation techniques. However, variations in loop gain, and thus bandwidth, can degrade the performance of fractional-N based transmit systems in which a fixed predistortion filter is used to compensate for the rolloff of the loop responses. Mismatch between the expected and actual loop response degrades the phase error of the transmitted signal. Simulations indicate that the loop gain must be accurate to within 15% of the expected nominal value for a 120 kHz loop bandwidth and fixed predistortion filter for less than 5 degrees rms phase error.
While the pole and zero locations, which depend on RC time constants, and the charge pump current, I
CP
, and loop divider variations can be compensated by methods known in the prior art, the VCO tuning gain, K
V
, poses a more difficult problem. The tuning gain, K
V
, of the VCO
18
characterizes the sensitivity of the VCO output frequency, F
VCO
, to changes in its tuning control voltage, V
CON
. The tuning gain, K
V
, is defined as:
K
v
≡
ⅆ
F
VCO
ⅆ
V
c
.
Eq
.
4
The tuning gain, K
V
, is usually not constant, and for an integrated, wide-range VCO, the tuning gain can vary as much as three to one over the desired tuning range.
Although methods of ‘flattening’ the variation in the tuning gain curve have been advanced, these methods typically require more complex tuning methods employing additional circuit elements. It is desirable to avoid any additional cost or complexity in the VCO, since the material cost, noise, and power consumption constraints are usually very tight.
Thus, instead of compensating the VCO
18
directly, the loop gain of the PLL
14
can be compensated by adjusting another gain term. For example, the charge pump current may be modified to compensate for variation in the tuning gain. This requires, first, a method of measuring the tuning gain, and second, a method of applying the appropriate adjustment.
Two methods have traditionally been used to characterize the tuning gain. The first is to measure or characterize the tuning gain of the VCO once and construct a table of tuning gain versus operating frequency, such a method is described in U.S. Pat. No. 4,893,087, issued Jan. 9, 1990, which is incorporated herein by reference. The table is then mapped to an adjustment factor for the charge pump current I
CP
versus the VCO divider value, and is stored in a non-volatile memory. While the nominal loop gain adjustments may be known for a ‘typical’ VCO
18
, the accuracy of nominal adjustments is not sufficient for high performance communications systems when integrated components with large tolerances are used. Hence, this method has the drawbacks of requiring an extensive one-time measurement process on each product that runs through the factory, and of requiring a non-volatile storage means within each product, which unduly increases costs.
The second method is to indirectly measure the closed loop bandwidth, which depends to a large extent on the loop gain. This type of method typically requires a modulation source to be applied to the VCO
18
, and a frequency discriminator to determine the frequency deviation of the VCO
18
while the PLL
14
is locked as noted in U.S. Pat. No. 5,079,522, issued Jan. 7, 1992, which is incorporated herein by reference. If the loop bandwidth is wider than the modulation frequency applied to the VCO
18
, the frequency deviation of the VCO
18
will be lower than if the loop bandwidth is narrower than
Nguyen Minh
RF Micro Devices, Inc.
Withrow & Terranova , PLLC
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