Wave transmission lines and networks – Coupling networks – With impedance matching
Reexamination Certificate
2002-06-27
2004-05-18
Le, Don (Department: 2819)
Wave transmission lines and networks
Coupling networks
With impedance matching
C333S124000, C333S032000
Reexamination Certificate
active
06737932
ABSTRACT:
BACKGROUND OF THE INVENTION
1. Statement of the Technical Field
The inventive arrangements relate generally to methods and apparatus for providing increased design flexibility for RF circuits, and more particularly for optimization of dielectric circuit board materials for improved performance.
2. Description of the Related Art
RF circuits, transmission lines and antenna elements are commonly manufactured on specially designed substrate boards. For the purposes of these types of circuits, it is important to maintain careful control over impedance characteristics. If the impedance of different parts of the circuit do not match, this can result in inefficient power transfer, unnecessary heating of components, and other problems. Electrical length of transmission lines and radiators in these circuits can also be a critical design factor.
Two critical factors affecting the performance of a substrate material are dielectric constant (sometimes called the relative permittivity or ∈
r
) and the loss tangent (sometimes referred to as the dissipation factor). The relative permittivity determines the speed of the signal in the substrate material, and therefore the electrical length of transmission lines and other components implemented on the substrate. The loss tangent characterizes the amount of loss that occurs for signals traversing the substrate material. Accordingly, low loss materials become even more important with increasing frequency, particularly when designing receiver front ends and low noise amplifier circuits.
Printed transmission lines, passive circuits and radiating elements used in RF circuits are typically formed in one of three ways. One configuration known as microstrip, places the signal line on a board surface and provides a second conductive layer, commonly referred to as a ground plane. A second type of configuration known as buried microstrip is similar except that the signal line is covered with a dielectric substrate material. In a third configuration known as stripline, the signal line is sandwiched between two electrically conductive (ground) planes. Ignoring loss, the characteristic impedance of a transmission line, such as stripline or microstrip, is equal to {square root over (L
l
/C
l
)} where L
l
is the inductance per unit length and C
l
is the capacitance per unit length. The values of L
l
and C
l
are generally determined by the physical geometry and spacing of the line structure as well as the permittivity of the dielectric material(s) used to separate the transmission line structures. Conventional substrate materials typically have a permeability of approximately 1.0.
In conventional RF design, a substrate material is selected that has a relative permittivity value suitable for the design. Once the substrate material is selected, the line characteristic impedance value is exclusively adjusted by controlling the line geometry and physical structure.
One problem encountered when designing microelectronic RF circuitry is the selection of a dielectric board substrate material that is optimized for all of the various passive components, radiating elements and transmission line circuits to be formed on the board. In particular, the geometry of certain circuit elements may be physically large or miniaturized due to the unique electrical or impedance characteristics required for such elements. For example, many circuit elements or tuned circuits may need to be an electrical ¼ wave. Similarly, the line widths required for exceptionally high or low characteristic impedance values can, in many instances, be too narrow or too wide for practical implementation for a given substrate. Since the physical size of the microstrip or stripline is inversely related to the relative permittivity of the dielectric material, the dimensions of a transmission line can be affected greatly by the choice of substrate board material.
From the foregoing, it can be seen that the constraints of a circuit board substrate having selected relative dielectric properties often results in design compromises that can negatively affect the electrical performance and/or physical characteristics of the overall circuit. An inherent problem with the conventional approach is that, at least with respect to conventional circuit board substrate, the only control variable for line impedance is the relative permittivity. This limitation highlights an important problem with conventional substrate materials, i.e. they fail to take advantage of the other factor that determines characteristic impedance, namely L
l
, the inductance per unit length of the transmission line.
A quarter-wavelength section of line can be designed to provide a match between a desired transmission line impedance and a given load. For example, in the circuit shown in
FIG. 1
, a transmission line can be matched to a load at the termination of the quarter-wave section if the characteristic impedance of the quarter wave section
Z
λ
4
is selected using the equation:
Z
λ
4
=
Z
01
Z
02
where
&AutoLeftMatch;
Z
λ
4
is
the
characteristic
impedance
of
the
quarter
-
wave
section
;
Z
01
is
the
characteristic
impedance
of
the
input
transmission
line
;
and
Z
02
is
the
load
impedance
.
Simple quarter-wave transformers will operate most effectively only over a relatively narrow bandwidth where the length of the transformer approximates a quarter-wavelength at the frequency of interest. In order to provide matching over a broader range of frequencies, a multi-section transformer can be designed with a large number of matching stages. For example, rather than attempting to use a single quarter-wave transmission line to transform from an impedance of 50 ohms to 10 ohms, one could use two quarter-wave sections in series. In that case, the first quarter wave section might be designed to transform from 50 ohms to 30 ohms, and the second quarter wave section might transform from 30 ohms to 10 ohms. Notably, the two quarter-wave sections when arranged in series would together comprise a half-wave section. However, this half wave section would advantageously function as a quarter-wave transformer section at half the design frequency. This technique can be used to achieve matching that is more broad-banded as compared to a simple quarter-wavelength section.
As the number of transformer stages is increased, the impedance change between sections becomes smaller. In fact, a transformer can be designed with essentially an infinite number of stages such that the result is a smooth, continuous variation in impedance represented in
FIG. 2
as Z(x) between feed line Z
0
and load Z
L
. In
FIG. 2
, x is the distance along the matching section. For maximally wide pass band response and a specified pass band ripple the taper profile has an analytic form known as the Klopfenstein taper. There is a substantial literature devoted to the design of multiple section and tapered transmission line transformers.
One problem with multiple transformer sections and tapered line transformers is that they are physically large structures. In fact, multiple section transformers are generally multi-quarter wavelengths long at the design frequency and tapered line transformers are generally at least about one wave-length long at the lowest design frequency and the minimum length is, to a degree, dependent on the impedance ratio. Accordingly, these designs are in many cases not compatible with the trend toward application of miniature semiconductors and integrated circuits.
Yet another problem with transmission line impedance transformers is the practical difficulties in implementation in microstrip or striplin
Killen William D.
Pike Randy T.
Harris Corporation
Le Don
Sacco & Associates PA
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