Communications: radio wave antennas – Antennas – High frequency type loops
Reexamination Certificate
1999-10-26
2001-12-11
Phan, Tho (Department: 2821)
Communications: radio wave antennas
Antennas
High frequency type loops
C343S741000, C343S792500, C343S793000, C343S807000
Reexamination Certificate
active
06329955
ABSTRACT:
FIELD OF THE INVENTION
The present invention relates generally to the field of broadband, reduced-size antennas for use in, e.g., HF and VHF communications, electromagnetic compatibility testing, electronic warfare, and ultrawideband and ground penetrating RADAR.
BACKGROUND OF THE INVENTION
For most applications, including both communications and electromagnetic compatibility testing, it is generally desirable for antennas to be as small as possible for reasons of convenience, durability, and aesthetics. In the case of military communications, it is also often necessary for antennas to exhibit low observability (LO). In the HF (3-30 MHz) and VHF (30-300 MHz) bands for which wavelengths are on the order of meters to tens of meters, it is thus necessary to utilize electrically-small antennas, that is, antennas with geometrical dimensions which are small compared to the wavelengths of the electromagnetic fields they radiate. Quantitatively, electrically-small antennas are generally defined as antennas which fit inside a so-called radiansphere, that is a sphere with a radius, r=&lgr;/2&pgr;, where &lgr; is the wavelength of the electromagnetic energy radiated.
Electrically-small antennas exhibit large radiation quality factors Q; that is, they store (on time average) much more energy than they radiate. This leads to input impedances which are predominantly reactive, and, as a result, the antennas can be impedance-matched only over narrow bandwidths. Furthermore, because of the large radiation quality factors, the presence of even small resistive losses leads to very low radiation efficiencies. According to known quantitative predictions of the limits on the radiation Q of electrically small antennas, the minimum attainable radiation Q for any linearly polarized antenna which fits inside a spherical volume of radius a can be computed exactly, according to the equation:
Q
=
1
ka
+
1
k
3
a
3
(Equ. 1)
where k=1/&lgr;, the wavenumber associated with the electromagnetic radiation. The available theories can be succinctly summarized by stating that the radiation Q of an electrically small antenna is roughly proportional to the inverse of its electrical volume. Furthermore, the radiation Q is essentially inversely proportional to the antenna bandwidth. Therefore, in order to achieve relatively broad bandwidth and high efficiency with a single-element electrically-small antenna of a given size, it is necessary to utilize as much as possible of the entire volume that an antenna occupies.
In order to achieve an antenna having this fundamental limit on radiation Q given in Equation 1, an antenna would have to excite only the TM
01
or TE
01
mode outside the enclosing spherical surface and store no electric or magnetic energy inside the spherical surface. So while, a Hertzian (short) dipole excites the TM
01
mode, it does not satisfy the criterion of storing no energy within the sphere and thus will exhibit a higher radiation Q (and hence narrower bandwidth) than that predicted by Equation 1.
In general, all antennas which radiate dipolar fields, such as wire dipoles and loops, are limited by the constraint given in Equation 1. Some broadband dipole designs have been successfully implemented and approach the limit given in Equation 1. However, it is not possible to construct a linearly-polarized, isotropic antenna which exhibits a radiation Q less than that predicted by Equation 1.
While Equation 1 represents the fundamental limit on the radiation Q for a linearly polarized, omni-directional antenna, it is not the global lower limit on radiation Q. Instead, an antenna which radiates equal power into the TM
01
and TE
01
modes can (in principle) achieve a radiation Q of:
Q
=
1
2
[
1
(
ka
)
3
+
2
ka
]
(Equ. 2)
A quality factor for an antenna which meets this characteristic is roughly half of that of an antenna which radiates only TM
01
or TE
01
, alone. As a result, the attainable impedance bandwidth of the antenna is nearly doubled. While an equipartition of radiated power in the two modes is required to achieve the radiation Q given in Equation 2, the polarization state and radiation pattern of the modes do not need to match, and instead can take on different forms depending on the relative phases and orientations of the modes. Although prior analysis has been performed on a very general class of antennas with equal electric and magnetic multipole moments, no specific antenna designs having these characteristics have been presented.
Ideal antennas having a pair of infinitesimally small, co-located, electric and magnetic dipoles oriented to provide orthogonal dipole moments have been theoretically and numerically examined previously and found to provide several useful features. Examples of such ideal antennas
10
,
20
are shown in
FIGS. 1 and 2
. The antennas
10
,
20
include an infinitesimal magnetic dipole loop
11
,
21
with an associated feed
12
,
22
and an infinitesimal electric (wire) dipole
13
,
23
with an associated feed
14
,
24
. As can be appreciated, because the antenna elements are infinitesimally small, the shape of the loop is not crucial. Thus, the square loop
21
in
FIG. 2
functions essentially equivalently to the circular loop
11
in FIG.
1
.
For the theoretically-examined co-located pair antenna described above, the electric field, in the far field region, is given by the equations:
E
θ
[
A
η
sin
θ
+
B
sin
φ
]
ⅇ
-
j
kr
r
,
and
(Equ. 3)
E
φ
=
B
[
cos
θcosφ
]
ⅇ
-
j
kr
r
(Equ. 4)
where A and B are weighting coefficients of the TM
01
and TE
11
modes respectively, and r, &thgr;, and &phgr; constitute a standard right-hand spherical coordinate system. If A=&eegr;B then the directional gain of the antenna is given by the equation:
G
(
θ
,
φ
)
=
3
[
(
sin
θ
+
sin
φ
)
2
+
cos
2
θcos
2
φ
]
4
(Equ. 5)
and cardioid patterns with linear polarization are provided in the &thgr;=90 plane and the &phgr;=90 plane.
FIG. 3
is a graph of the farfield gain pattern. As can be seen, a maximum gain of G
max
=3.0 (4.77 dBi) is achieved at &thgr;=90 and &phgr;=90.
However, if A=j&eegr;B, the directional gain is:
G
(
θ
,
φ
)
=
3
[
(
sin
θ
)
2
+
(
sin
φ
)
2
]
4
(Equ. 6)
The farfield gain pattern of such an antenna is depicted in FIG.
4
. The maximum gain still occurs at &thgr;=90. However, for this configuration, the maximum gain value
Gmax
is now only 1.5 (1.77 dBi). Therefore, as can be appreciated by one of skill in the art, the combination of an electric and a magnetic dipole with proper orientation, amplitude ratio, and relative phase results in a radiator with roughly half the radiation Q and as much as 3 dB more gain than an isolated dipole.
Another useful aspect of including both electric and magnetic dipole modes in an antenna is that the maximum power output (as limited by electric field breakdown in the nearfield) is improved. It can be seen physically and has been shown mathmatically, that, for purposes of producing maximum radiated power before electric field breakdown in the nearfield, the TE (magnetic multipole) modes and in particular, the TE
01
mode are better. This is because the nearfield energy is magnetic as opposed to electric. Thus any admixture of TE modes is an improvement over a simple dipole antenna.
Previous work in this area has been limited primarily to theoretical and numerical investigations of co-located pairs of infinitesimal electric and magnetic dipoles (as well as ensembles of higher-order multipoles). While, as discussed above, the co-located pair of infinitesimal electric and magnetic dipoles has been shown to possess many valuable attributes, it is not a practical radiator. First, co-location is impossible w
Crook Gentry Elizabeth
McLean James Stuart
Darby & Darby
Phan Tho
TDK RF Solutions Inc.
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