Wave transmission lines and networks – Plural channel systems – Having branched circuits
Reexamination Certificate
2001-12-27
2004-11-02
Pascal, Robert (Department: 2817)
Wave transmission lines and networks
Plural channel systems
Having branched circuits
C333S134000, C333S202000, C333S235000
Reexamination Certificate
active
06812808
ABSTRACT:
FIELD OF THE INVENTION
The invention is related to the field of combiners. More particularly, this invention relates to inline combiner networks which combine multiple frequency sources.
BACKGROUND OF THE INVENTION
FIGS. 1 and 2
illustrate a combining network having two cavity resonators which uses intrusive coupling loops to couple signals from the different resonators. This approach has been used with ceramic, waveguide, and coaxial resonators. Coupling of a signal from each cavity is achieved in the following manner. A loop is placed into the cavity such that it couples into the magnetic field of the desired mode. The two loops (one for each cavity) are then joined at a common terminal and connected to the antenna port.
FIG. 3
shows a schematic of a general two-channel cavity combiner. The resonators are treated as a parallel LC resonator that is mutually coupled to two ports. The input port is connected—usually through an isolator—to a transmitter. The output port is connected to a junction via a transmission line, and a shunt component is attached at the junction to remove excess inductive reactance.
The resonator itself is used to pass the primary frequency while rejecting other frequencies by a certain amount.
The frequency response of a cavity centered at a frequency f
0
is given in equation 1:
H
(
f
)
=
(
1
-
Q
L
Q
U
)
·
1
1
+
(
2
·
Q
L
·
f
-
f
0
f
0
)
2
(
1
)
where Q
L
is the ratio of the center frequency of the resonator to the frequency separation between the half-power (3 dB) points and is a function of the cavity coupling. The term Q
U
is the unloaded Q of the resonator and represents the resonator Q if there was no external loading. The ratio of loaded Q to unloaded Q is the reflection coefficient at the center frequency of the resonator due to the internal losses of the resonator. The closer the ratio is to unity, the higher the loss in the cavity at midband. An important tradeoff in cavity performance is between narrow bandwidth and low loss.
The electrical length of the lines separating the resonators from the junction is determined from transmission-line theory. In transmission-line theory, it is widely known that an ideal line of length L transforms a load whose admittance is Y to an admittance Y
B
such that:
Y
B
=
Y
0
·
(
cos
(
2
·
π
·
L
λ
)
·
Y
+
1
i
·
sin
(
2
·
π
·
L
λ
)
·
Y
0
)
(
cos
(
2
·
π
·
L
λ
)
·
Y
0
+
1
i
·
sin
(
2
·
π
·
L
λ
)
·
Y
)
(
2
)
where Y
0
is the characteristic admittance of the transmission line, and &lgr; is the wavelength in the transmission line. This equation is found as equation 14 in Ramo, S; Whinnery, J.; Van Duzer, T.; Fields and Waves in Communications Electronics, 3
rd
Edition., 1994, John Wiley & Sons, New York, pp229-232, p254-256, hereby incorporated by reference. The transmission line can be several different shapes, such as coaxial or parallel wire. The embodiment we use uses a air-dielectric microstrip line designed such that the characteristic impedance Z
0
is 50 ohms, which corresponds to a characteristic admittance Y
0
of 1/Z
0
or 0.02 mhos.
One of the well known property of ideal transmission lines is that the impedances tend to repeat themselves every half-wavelength. For example, a shorted transmission line (Y→∞) acts like an open circuit when the distance from the short is &lgr;/4—one quarter wavelength. When the distance reaches &lgr;/2—one half wavelength—the admittance is that of short-circuit again. The impedance curves can be found in Pozar, D.; Microwave Engineering, 1993, Addison Wesley, New York, pp 76-84, hereby incorporated by reference. In the case where the admittance is Y, the transformed admittance Y
B
is given in equation 3.
Y
B
=
Y
0
2
Y
(
3
)
Equation 3 shows that the quarter-wave transmission line acts as an admittance inverter because the higher admittances become low admittances at the opposite end of the transmission line.
The admittance of the isolated resonator loaded on the output with a load with admittance Y
0
is approximately given as equation 4.
Y
=
Y
0
·
(
1
+
Q
L
Q
U
)
·
(
1
+
2
j
·
Q
L
·
f
-
f
0
f
0
)
(
4
)
Equation 4 shows that the admittance Y becomes very large as the frequency f becomes more distant from f
0
. This means that an ideal parallel resonator becomes a short circuit at frequencies far from resonance, and a quarter-wave resonator will transform the near-short circuit.
Using the preferred embodiment as shown in
FIG. 3
, the resonators are set for center frequencies of f
1
for the TX
1
cavity and f
2
for the TX
2
cavity. In an ideal parallel-cavity resonator, the electrical length of the loop would be zero, and the cavity resonator's off-resonance admittance would approach the infinite conductivity of a short circuit as the TX
2
resonator frequency becomes further from f
2
. In such a case, attaching a transmission line of a quarter-wavelength would make the cavity look like a very low admittance and approach an open-circuit off the resonant frequency of the cavity at the other end of the cable.
If this admittance was placed in parallel with the antenna which is assumed to have an admittance of Y
0
, then the additional “shunting” loss &agr;
sh
caused by the joined cavity is given in equation 5.
α
s
h
=
&LeftBracketingBar;
2
2
+
Y
B
Y
0
&RightBracketingBar;
(
5
)
As the magnitude of the Y
B
/Y
0
ratio approaches zero, the shunting loss approaches zero. This is expected since an open circuit in parallel with any admittance has no effect on said admittance. If a second cavity on a frequency sufficiently separated from the first cavity is also attached to a quarter-wave transmission line, they can be joined to a common output. The first cavity on its resonant frequency only sees a small additional loading from the second cavity and vice versa.
As equation 4 shows, the cavity's frequency response has an effect on the admittance off resonance or off the cavity's resonant frequency. However, the combiner can still be used to combine cavities as long as the frequency separation between cavities is such that the response of one cavity frequency on the neighbor's cavity response is down 4-6 dB from the center of the response. In such a case, the shunting loss approaches 1.3 dB. The shunting loss can be as high as 1.5 dB with multiple channels and still be useable in most systems where frequency separations are tight.
Ideally, the two loops in
FIGS. 1 and 2
should be separated electrically from the junction by a transmission line whose length is one-quarter of a wavelength. In such a case, the shunt reactance shown in
FIGS. 3 and 4
would be unnecessary. Unfortunately, an exact quarter-wave line is difficult to define or achieve. For example, all cavities have some small inductive reactance due to the finite length of the loop.
FIG. 3
shows the general case where the line separating the cavities in the combiner is less than—but fairly close to—a one-quarter-wavelength transmission line. The schematic includes the inductive reactance of the loop. Though not an exact quarter-wave line length, the two resonators can be connected as shown as long as the internal shunt reactance at the junction is cancelled using a shunt network. In the case where the separating lines are less than a quarter-wave in length, the internal shunt reactance at the junction is cancelled using a capacitor C
bal
is shown in FIG.
3
.
The main difficulty with using internal loops to couple signals from the cavity resonator is the electrical length required to reach the strong field region—particularly in ceramic resonators. Because of the cavity size, the loop become so long that the lines are longer than quarter-wave. In the case where the lines are longer than a quarter-wavelength but less than a multiple of a half-wavelength, a sh
Gaukel Kevin M.
Kulaga Thomas J.
Pascal Robert
Radio Frequency Systems Inc.
Takaoka Dean
LandOfFree
Aperture coupled output network for ceramic and waveguide... does not yet have a rating. At this time, there are no reviews or comments for this patent.
If you have personal experience with Aperture coupled output network for ceramic and waveguide..., we encourage you to share that experience with our LandOfFree.com community. Your opinion is very important and Aperture coupled output network for ceramic and waveguide... will most certainly appreciate the feedback.
Profile ID: LFUS-PAI-O-3337367