Analog discrete-time FIR filter

Details Analog discrete-time FIR filter Analog discrete-time FIR filter

2000-09-20

2003-03-04

Mai, Tan V. (Department: 2124)

Electrical computers: arithmetic processing and calculating

Electrical digital calculating computer

Particular function performed

C708S319000, C708S819000

Reexamination Certificate

active

06529926

ABSTRACT:

BACKGROUND OF THE INVENTION
For signals with very high bandwidth, analog filtering is often more economical than digital filtering. Even where costs are less important, there are still cases where analog filtering is the only viable method for performing signal-processing functions.
Examples of high-bandwidth signals that may require analog filtering include disk drive read channels, radio communication channels, wireline communication channels and fiber-optic communication channels. The types of processing that must be performed on these signals include channel response equalization and channel phase compensation.
Three broad categories encompass several prior art analog filters. These categories may be distinguished based on the type of method each uses to create signal delays during the filtering process.
In
FIG. 1
, a representative continuous-time integrator based filter is shown and generally designated
100
. Filter
100
, and other filters included in this category generate signal delays using a series of integrators
102
(that are individually labeled in
FIG. 1
as
102
a
-
102
n
). Each integrator
102
is typically implemented as an inductor or capacitor. The primary disadvantage of this type of filter is that it cannot compensate for distributed or noncausal errors in the channel.
FIG. 2
shows an example of a continuous-time transmission-line based filter
200
. Unlike Filter
100
, Filter
200
has a finite impulse response (FIR) (i.e., the filter is non-recursive and does not process feedback). For Filter
200
, signal delays are generated by a plurality of transmission lines
202
. Each transmission line
202
C is typically implemented as a stripline. The primary disadvantage of the type of filter is that the transmission lines are physically large. As a result, it is difficult to implement circuits of this type in integrated-circuit technologies.
FIG. 3
shows an example of a discrete-time analog filter
300
. In this type of filter, delays are generated by a series of sample-and-hold circuits
302
. Filter
300
has a finite impulse response (FIR) and is further described in U.S. Pat. No. 4,316,258, issued to Berger, et. al., for an invention entitled “Digitally programmable filter using electrical charge transfer”.
Operation of filter
300
can be described as follows: The first sample and hold circuit
302
a
samples the input signal x(t) at uniformly spaced times 0, T, 2T, . . . and generating samples x(
0
), x(T), x(
2
T), . . . The second sample and hold circuit
302
b
samples the output of sample and hold circuit
302
a
before sample and hold circuit
302
a
acquires a new sample (e.g., between times O and T), thus obtaining the previous sample value (e.g., x(o)). Therefore, at time t=kT, the output of circuit
302
a
is x(kT) whereas the output of circuit
302
b
is x((k−1)T). Each sample and hold circuit
302
I behaves in this manner relative to its preceding circuit. The output of each sample and hold circuit
302
n
at time t=kT is therefore as shown in FIG.
3
. To perform the filtering, each of the output samples Sn from each circuit
302
n
is multiplied by a coefficient Cn, and the resulting products are then added together. The result is a filtered version of the sampled input signal with the filter transfer function given by:
H
(
z
)=
C
0
+C
1
z
−1
+C
2
z
−2
+ . . .
where z is the unit delay operator. The filter architecture of
FIG. 3
suffers from several significant disadvantages. First each sample and hold circuit
302
n
samples during the hold phase of the preceding sample and hold circuit
302
in the pipeline, thus requiring two track-and-holds for each filter tap. Second, noise, offset, and nonlinearity errors accumulate as the signal propagates along the chain of sample and hold circuits
302
. See S. Kiriaki, T. L. Viswanathan, G. Feygin, B. Stazewski, R. Pierson, B. Krenik, M. de Wit, K. Nagaraj, “A 160-MHz Analog Equalizer for Magnetic Disk Read Channels”, IEEE Journal of Solid State Circuits, vol. 32, no. 11, November 1997, pp. 1839-1850.
An architecture that does not have these two disadvantages is illustrated by filter
400
of FIG.
4
. Filter
400
has a finite impulse response. Filter
400
includes a series of n+1 track and hold circuits
402
, a crosspoint switch matrix
404
, a series of n multipliers
406
and an adder
408
.
Track and hold circuits
402
have two operational states. During track mode, track and hold
402
transfers their input to their outputs with a gain of one. During hold mode, track and hold circuits
402
outputs their last transferred values. This differs from sample and hold circuits, which output sample value without having a tracking (transfer or pass-through) mode.
Crosspoint switch matrix
404
has n+1 inputs and n outputs. Each track and hold circuit
402
is connected to one of these inputs. Each of these outputs is connected to a respective multiplier
406
i
. The outputs of multipliers
406
are connected to the n inputs of an adder
408
. Crosspoint
404
switch matrix contains a switch connecting each of its (n+1) inputs to each of its n outputs, with one switch per output being closed at any given time. This allows crosspoint switch matrix
404
to select any set of n inputs from among the n+1 inputs and pass that set of n inputs to its n outputs.
A control circuit (not shown) clocks and controls the operation of filter
400
. During each clock period, the control circuit causes one track and hold circuit
402
(known as the active track and hold circuit
402
) to track the input signal. This means that the active track and hold
402
transfers its input (the input signal) to its output with a gain of one. The control circuit causes the remaining track and hold circuits
402
to remain in hold mode. Each of these track and hold circuits
402
outputs its last transferred value of the input signal. During subsequent clock periods, the control circuit causes the active track and hold circuit
402
to rotate among the series of track and hold circuits
402
.
At time t=kT, valid samples will be present in the inactive track and holds, the samples representing x((k−1)T), x((k−2)T), . . . x((k−n)T). The location of the samples at the inputs to matrix
404
will be different at each instant of time. However, because of the rotating nature of the sampling, e.g., the control circuit configures the crosspoint switch matrix
404
to map the inputs from the appropriate track and hold circuits
402
(i.e., the track and hold circuits
402
having valid sample values) to respective multipliers
406
. The multiplied samples are forwarded to adder
408
. Adder
408
sums the multiplied samples to form a filtered output signal y[k].
Filter
400
suffers from several disadvantages. First, the number of switches in crosspoint switch matrix
404
grows roughly as the square of the number of taps:
N
switch
=n
(
n+
1).
The large number of switches results in a large parasitic capacitance at each of the input and output terminals of crosspoint switch matrix
404
. This limits the speed of operation of the circuit. Second, the sampled signal must traverse the entire signal path, including crosspoint switch matrix
404
, multipliers
406
and adder
408
, within one clock cycle. For systems with high sampling rates (typically above 1-5 GHz), certain integrated circuit technology is not fast enough to perform all the processing with sufficient accuracy within the sample period. As a result, filters using this architecture may suffer from a bottleneck in terms of sampling rate and accuracy.
Additional description of Filter
400
may be found in: 1) S. Kiriaki, T. L. Viswanathan, G. Feygin, B. Stazewski, R. Pierson, B. Krenik, M. de Wit, K. Nagaraj, “A 160-MHz Analog Equalizer for Magnetic Disk Read Channels”, IEEE Journal of Solid State Circuits, vol. 32, no. 11, November 1997, pp. 1839-1850. 2) Kiriaki, et al., “FIR filter architecture”, U.S. Pat. No. 6,035,320, Mar. 7, 2000. 3) Carley,

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