Oscillators – Solid state active element oscillator – Transistors
Utility Patent
1999-05-19
2001-01-02
Mis, David (Department: 2817)
Oscillators
Solid state active element oscillator
Transistors
C331S1160FE, C331S138000, C331S139000
Utility Patent
active
06169459
ABSTRACT:
CROSS REFERENCE TO RELATED APPLICATIONS
(Not Applicable)
BACKGROUND OF THE INVENTION
The oscillator is one of the basic electronic circuits, and typically includes a semiconductor amplifier powered by a DC voltage and which has a feedback loop between the amplifier output and input to cause an AC output. There are many trade-offs for the oscillator designer, who must balance factors such as size, power requirements, and ease of manufacture with performance requirements such as waveform shape, amplitude stability and frequency stability.
The bridge oscillator is a known device having four impedances connected in a ring to form a four terminal bridge, with the two inputs of a differential amplifier connected to opposing terminals and the output of the amplifier and ground connected to the other opposing terminals. If a piezoelectric crystal is utilized as one of the impedances, the combination of positive and negative feedback loops can provide high circuit Q and thus produce very stable oscillation.
The standard bridge oscillator (SBO), and variations thereon, is discussed in some detail in U.S. Pat. No. 4,661,785 of Benjaminson.
FIG. 1
of the Benjaminson patent shows a generic standard is recreated as
FIG. 1
herein. The loop-gain equation for this oscillator is:
A=(&bgr;
p
−&bgr;
n
)A
v
=1∠0° (1)
where A is the loop-gain and A
v
is the gain of the amplifier. For oscillation A must equal 1. Also, positive feedback &bgr;
p
must be greater than negative feedback &bgr;
n
for the phase condition of oscillation to be met.
For the SBO of FIG.
1
,
&bgr;
p
=R
2
/(R
2
+R
1
) and &bgr;
n
=Z
res
/(Z
res
+R
f
) (1a)
where Z
res
, the impedance of crystal resonator Y
1
, is:
Z
res
=
R
m
+
j
L
m
ω
+
1
j
C
m
ω
=
R
m
+
j
ω
2
L
m
C
m
-
1
ω
C
m
(
2
)
where R
m
, C
m
, and L
m
are the motional parameters of the quartz resonator. Eq. (2) assumes that R
m
is much smaller than the reactive impedance of the resonator shunt capacitance, C
0
. Using this assumption, Z
res
accurately describes the resonator impedance at frequencies near series resonance.
Since the SBO is a series resonant oscillator, Z
res
at series resonance is equal to R
m
, the motional resistance of the resonator. Hence the negative feedback term at series resonance is:
&bgr;
n
=R
m
/(R
m
+R
f
) (3)
If there is excess loop-gain and the loop-phase is zero, then the nonlinear properties of the gain stage will limit A
v
, to a value commensurate with eq. (1).
The term (&bgr;
p
−&bgr;
n
), as determined from eqs. (1a), contains all the frequency-dependent loop-phase information, as &bgr;
n
has a phase versus frequency function dependent upon the values of R
f
and Z
res
. This frequency dependent vector is subtracted from a fixed real quantity, &bgr;
p
. The resulting vector, (⊖
p
−&bgr;
n
), determines the oscillator loop-phase. As the magnitude of &bgr;
p
approaches &bgr;
n
, with &bgr;
p
>&bgr;
n
, d&thgr;/d&ohgr; of the resulting vector is greater than d&thgr;/d&ohgr; of &bgr;
n
. The penalty paid for this increased phase-rate vector is that the magnitude of this resulting vector decreases, requiring larger A
v
to sustain oscillation. Also, as the resulting vector becomes small, it is more prone to 1/f noise in the sustaining amplifier.
The circuit loop-phase shift relative to frequency (&ohgr;) is adjustable by choosing the feedback ratios, &bgr;
p
and &bgr;
n
, such that the desired loop Q is achieved. To determine the effect &bgr;
p
and &bgr;
n
have on d&thgr;/d&ohgr; of the loop-phase, an equation for the loop-phase versus &ohgr; is required. The loop-phase d&thgr;/d&ohgr; will be compared to the d&thgr;/d&ohgr; of resonator impedance (Z
res
) phase to determine the relative loop Q. Let M
Q
be the ratio of the derivatives of oscillator loop-phase to resonator impedance phase with respect to &ohgr;. M
Q
is a measure of the loaded Q of the oscillator in that Q can be equated to the derivative of phase change relative to &ohgr;.
After an algebra exercise which is not necessary for an understanding of the invention, the expression for the magnitude of M
Q
is determined to be:
M
Q
=
(
ⅆ
(
θ
Loop
)
/
ⅆ
ω
)
/
(
ⅆ
(
θ
Resonator
)
/
ⅆ
ω
)
=
&LeftBracketingBar;
(
β
n
-
1
)
(
β
n
)
β
p
-
β
n
&RightBracketingBar;
(
4
)
The aforementioned Benjaminson patent discloses a different value for M
Q
in eq. (27):
M
Q
=
β
n
β
p
-
β
n
(
Benjaminson
)
(
5
)
Eq. (5) was derived using the assumption that &bgr;
n
is proportional to R
m
, which assumption is true only when &bgr;
n
<<1. These facts explain and help verify the observation that eq. (4) is a general case solution for the bridge oscillator described in FIG.
1
. These results have been verified with SPICE analysis and negative resistance modeling. For any further reference to M
Q
, eq. (4) is the assumed definition.
Practical Realization of the SBO
FIG. 5
of the Benjaminson patent discloses a practical realization of an SBO, and is recreated as
FIG. 2
herein. This SBO has some inherent shortcomings and does not well represent the ideal bridge oscillator depicted in the “block” schematic, FIG.
1
. First, this design uses a high-impedance collector node (of 302) to drive the bridge network, although &bgr;
n
of the bridge may be a relatively low impedance. Another disadvantage with the SBO is that only the differential amplifier nonlinearities, h, control the amplitude of oscillation. As a transistor base-emitter ac voltage increases, so does the effective nonlinear value of intrinsic base-emitter impedance h=1/g
m
, where g
m
is the transconductance of the device. For high M
Q
operation, the signals at each base are approximately the same amplitude. This means that the ac base-to-emitter voltages of the differential amplifier transistors are relatively small, even for large signals at each base. Therefore, large changes in amplitude result in only small changes in operating h. This means that the oscillation amplitude will vary greatly with respect to resonator resistance, temperature, etc. Therefore this circuit must be used with an automatic level control (ALC) circuit 320 which controls some element in the oscillator (bias or feedback component) continuously to maintain a desired signal amplitude.
An additional disadvantage is that the bridge oscillator typically utilizes a number of large value capacitors and inductors (such as the resonant circuit of CF and LF, and RFC 304) which makes it difficult to implement as an integrated circuit. For these and other reasons, bridge oscillators are typically not used in mainstream designs; rather, they are generally relegated to relatively low-volume, high-performance applications.
SUMMARY OF THE INVENTION
The active bridge oscillator (ABO) of this invention is a bridge-type oscillator design that is easy to design and overcomes many of the operational and design difficulties associated with standard bridge oscillator (SBO) designs. The ABO will oscillate with a very stable output amplitude over a wide range of operating conditions without the use of an automatic-level-control (ALC). It is also designed for implementation as an integrated circuit and can be designed for use over a very wide frequency range with only minor circuit variations. The ABO is a practical high-performance alternative to the SBO that delivers most of the performance advantages of a bridge oscillator without the inherent disadvantages of complexity, design difficulty, high-power requirements, and low integration potential.
To achieve the foregoing and other objects, and in accordance with the purpose of the present invention, as embodied and broadly described herein, the present invention may comprise an active bridge oscillator including a differential amplifier having an inverting input on a first gain device and a noninverting input on a s
Libman George H.
Mis David
Sandia Corporation
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