Phase shifter, attenuator, and nonlinear signal generator

Details Phase shifter, attenuator, and nonlinear signal generator Phase shifter, attenuator, and nonlinear signal generator

2000-03-31

2003-02-18

Pascal, Robert (Department: 2817)

Wave transmission lines and networks

Coupling networks

Delay lines including long line elements

C333S118000

Reexamination Certificate

active

06522221

ABSTRACT:

BACKGROUND OF THE INVENTION
The present invention relates to a small phase shifter, attenuator, and nonlinear signal generator having matched input and output impedances.
With the recent rapid progress of wireless multimedia communication, demands for smaller and more economical wireless devices are increasing. A monolithic microwave integrated circuit (MMIC) has attracted attention as a basic technology for advancing the miniaturization and economization of wireless devices for the following reasons. That is, not only the MMIC itself is small, but also the mass-productivity increases because highly uniform chips can be fabricated with no adjustment by a semiconductor process. Furthermore, high-degree integration and high-accuracy reproduction can reduce the packaging cost and improve the reliability.
Known examples of high-frequency functional circuits expected to be miniaturized by the MMIC are an amplifier for amplifying a high-frequency signal, an oscillator for generating a local oscillation signal, and a frequency converter for performing frequency conversion. Additionally, for the purpose of applying to an antenna directivity control circuit or a distortion compensation circuit of a power amplifier, it is also being expected to miniaturize, by the MMIC, a phase shifter for controlling the phase of a high-frequency signal, an attenuator for attenuating the amplitude of a high-frequency signal, and a nonlinear signal generator for generating a nonlinear signal.
A conventional phase shifter and attenuator will be described below.
FIG. 62
shows the conventional phase shifter and attenuator. These phase shifter and attenuator are a reflection-type phase shifter and attenuator using a 90° branch line hybrid. The basic operating principle of this phase shifter is described in, e.g., [7.2 Analogue implementations, pp. 261-265, I. D. Robertson, “MMIC Design,” London, IEE, 1995] and [11.6 Varactor Analogue Phase Shifter, pp. 193-195, J. Helszajn, “Passive and active microwave circuits,” New York, John Wiley & Sons, 1978]. Also, the basic operating principle of this attenuator is described in [8.5.1 Analogue reflection-type attenuator, pp. 332-333, I. D. Robertson, “MMIQ Design,” London, IEE, 1995].
As shown in
FIG. 62
, the 90° branch line hybrid is composed of four high-frequency transmission lines
3
a,
3
b,
3
c,
and
3
d
whose electrical length at frequency f
0
is 90°. The connecting nodes of these high-frequency transmission lines
3
a
to
3
d
are I/O terminals
4
a,
4
b,
4
c,
and
4
d
of the 90° branch line hybrid. An input port
1
is connected to the I/O terminal
4
a
of the 90° branch line hybrid. An output port
2
is connected to the I/O terminal
4
b
of the 90° branch line hybrid. Also, variable impedance elements
5
a
and
5
b
are connected to the I/O terminals
4
c
and
4
d,
respectively, of the 90° branch line hybrid.
Let Z
0
be the input and output impedances of the input and output ports
1
and
2
, Z
0
be the characteristic impedance of the high-frequency transmission lines
3
a
and
3
b,
Z
0
/{square root over ( )}2 be the characteristic impedance of the high-frequency transmission lines
3
c
and
3
d,
and Z
1
be the impedance of the variable impedance elements
5
a
and
5
b.
The operation of the conventional arrangement shown in
FIG. 62
will be described below. An input signal from the input port
1
is distributed by the 90° branch line hybrid constituted by the high-frequency transmission lines
3
a
to
3
d
and output from the I/O terminals
4
c
and
4
d
of this 90° branch line hybrid. These I/O terminals
4
c
and
4
d
are terminated by the variable impedance elements
5
a
and
5
b,
respectively. Therefore, a portion of the signal power is absorbed by a resistance component R
1
of the impedance Z
1
, and the rest of the signal is given a phase change by a reactance component X
1
of the impedance Z
1
and reflected to the input port
1
and the output port
2
.
Since the variable impedance elements
5
a
and
5
b
have the same impedance Z
1
, the signals reflected from the variable impedance elements
5
a
and
5
b
to the input port
1
have equal amplitudes and opposite phases and thereby cancel each other out. The signals reflected from the variable impedance elements
5
a
and
5
b
to the output port
2
are synthesized with equal amplitudes and the same phase. Accordingly, by changing the impedance Z
1
of the variable impedance elements
5
a
and
5
b,
it is possible to allow the configuration shown in
FIG. 62
to operate as a phase shifter or an attenuator while keeping the I/O impedance matching at the frequency f
0
.
To allow the configuration shown in
FIG. 62
to operate as a phase shifter, it is only necessary to set the variable impedance elements
5
a
and
5
b
such that the impedance Z
1
is substantially constituted by the reactance component X
1
, and continuously change this reactance component X
1
. A phase change amount &thgr; of the phase shifter when the reactance component is changed from X
1
to (X
1
+&Dgr;X
1
) is given by
θ
=
-
2
⁢
tan
-
1
⁡
(
X
1
+
Δ
⁢
 
⁢
X
1
Z
0
)
+
2
⁢
tan
-
1
⁡
(
X
1
Z
0
)
⁡
[
rad
]
(
1
)
To permit the configuration shown in
FIG. 62
to operate as an attenuator, it is only necessary to set the variable impedance elements
5
a
and
5
b
such that the impedance Z
1
is substantially constituted by the resistance component R
1
, and continuously change this resistance component R
1
. An attenuation amount L of this attenuator is given by
L
=
20
⁢
log
10
⁢
&LeftBracketingBar;
Z
0
+
R
1
Z
0
-
R
1
&RightBracketingBar;
⁡
[
dB
]
(
2
)
FIG. 63
shows a practical example of the conventional phase shifter shown in FIG.
62
. The same reference numerals as in
FIG. 62
denote the same parts in
FIG. 63
, and a detailed description thereof will be omitted. This phase shifter shown in
FIG. 63
uses variable capacitors
11
a
and
11
b
as the variable impedance elements
5
a
and
5
b,
respectively. Assume that the high-frequency transmission lines
3
a
to
3
d
are lossless, the I/O impedance Z
0
=50&OHgr;, and the frequency f
0
=5 GHz.
FIG. 64
shows the simulation results of the amplitude characteristics (a forward transfer factor S
21
and an input reflection coefficient S
11
). The abscissa indicates the frequency [GHz], the left ordinate indicates the forward transfer factor S
21
[dB], and the right ordinate indicates the input reflection coefficient S
11
[dB].
FIG. 65
shows the simulation results of the phase characteristic (forward transfer factor S
21
). The abscissa indicates the frequency [GHz], and the ordinate indicates the forward transfer factor S
21
[deg.] Referring to
FIGS. 64 and 65
, a capacitance C
1
of the variable capacitors
11
a
and
11
b
is changed to 0.05, 0.1, 0.3, 0.5, and 0.7 pF. As shown in
FIGS. 64 and 65
, at frequency f=4.5 GHz to 5.4 GHz, an amplitude fluctuation is 0.5 dB or less, an input reflection amount is −10 dB or less (FIG.
64
), and a phase change amount is 60° or more (FIG.
65
).
FIG. 66
shows a practical example of the conventional attenuator shown in FIG.
62
. The same reference numerals as in
FIG. 62
denote the same parts in
FIG. 66
, and a detailed description thereof will be omitted. The attenuator shown in
FIG. 66
uses variable resistors
21
a
and
21
b
as the variable impedance elements
5
a
and
5
b,
respectively. Assuming that the high-frequency transmission lines are lossless, the I/O impedance Z
0
=50&OHgr;, and the frequency f
0
=5 GHz.
FIG. 67
shows the simulation results of the amplitude characteristic (forward transfer factor S
21
). The abscissa indicates the frequency [GHz], and the ordinate indicates the forward transfer factor S
21
[dB].
FIG. 68
shows the simulation results of the amplitude characteristic (input reflection coefficient S
11
). The abscissa indicates the frequency [GHz],

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