Soft-switched full-bridge converter

Details Soft-switched full-bridge converter Soft-switched full-bridge converter

2000-08-31

2002-05-21

Han, Jessica (Department: 2838)

Electric power conversion systems

Current conversion

Including d.c.-a.c.-d.c. converter

C363S056020, C363S132000

Reexamination Certificate

active

06392902

ABSTRACT:

BACKGROUND OF THE INVENTION
1. Field of the Invention
This invention relates to an isolated dc/dc converter. More particularly, this invention relates to a constant-frequency, isolated dc/dc full-bridge converter that operates with zero-voltage switching of the primary switches.
2. Discussion of the Related Art
A factor adversely affecting the performance of a conventional “hard-switched” pulse-width-modulated (PWM) converter at a high switching frequency is circuit parasitics, such as semiconductor junction capacitances, transformer leakage inductances, and rectifier reverse recovery. Generally, these parasitics introduce additional switching losses and increase component stresses, thus limiting the converter's maximum operation frequency. To operate a converter at a high switching frequency and to achieve a high power density, elimination or a reduction of circuit parasitics without degrading conversion efficiency is required. One approach which incorporates circuit parasitics into circuit operations uses a resonant technique or a constant-frequency PWM soft-switching technique.
Under a resonant technique, a resonant tank circuit shapes the current or voltage waveforms of semiconductor switches in the converter to create either zero-current turn-off, or zero-voltage turn-on conditions. However, relative to conventional switching techniques, zero-current switching (ZCS) and zero-voltage switching (ZVS) in a resonant-type converter cause higher current or voltage stresses in the semiconductor switches. In addition, to create a ZCS or a ZVS condition, a resonant topology typically circulates a significant amount of energy. Thus, the trade-off between switching loss and conduction loss may result in a lower efficiency or a larger high-frequency resonant-type converter, when compared to a PWM counterpart operating at a lower frequency, especially in an application involving a wide input voltage range. In addition, variable frequency operation is often seen as a disadvantage of resonant converters. As a result, although resonant converters are used in a number of niche applications, such as those with pronounced parasitics, the resonant technique has not gained wide acceptance in the power supply industry in high-frequency, high-power-density applications.
To overcome the degradation of efficieny due to circuit parasitics, a number of techniques that enable a constant-frequency PWM converter to operate with ZVS or ZCS have been proposed. In such a soft-switching PWM converter—one that possess the PWM-like square current and voltage waveforms—lossless turn-off or turn-on of the switches can be achieved without a significant increase of conduction loss. One soft-switched PWM circuit is the soft-switched, full-bridge (FB) PWM converter 100 of FIG.
1
(
a
), which is discussed in the article “Pseudo-Resonant Full Bridge DC/DC Converter,” by O. D. Petterson, D. M. Divan, published in
IEEE Power Electronics Specialists' Conf: Rec.,
pp. 424-430, 1987, and in the article “Design Considerations for High-Voltage High-Power Full-Bridge Zero-Voltage-switched PWM Converter,” by J. Sabate, et. al., published in
IEEE Applied Power Electronics Conf:
(
APEC
)
Proc.,
pp. 275-284, 1990. Converter 100 provides ZVS in the primary switches with relatively small circulating energy and at a constant switching frequency. A constant-frequency output voltage is achieved by a phase-shift technique. Under this technique, a switch in the lagging leg (i.e., switches
103
and
104
) of the bridge is closed only after a delay (i.e., phase shifted) relative to the closing of a corresponding switch in the leading leg (i.e., switches
101
and
102
), as shown in FIG.
1
(
b
). Without the phase-shift, no voltage is applied across the primary winding
105
a
of transformer
105
, resulting in a zero output voltage. However, if the phase-shift is 180°, the maximum volt-second product is applied across the primary winding
105
a
, which produces a maximum output voltage. In converter
100
of FIG.
1
(
a
), a ZVS condition in the lagging-leg (i.e., switches
103
and
104
) is achieved by the energy stored in output filter inductor
106
. Since filter inductor
106
is relatively large, the energy stored in filter inductor
106
is sufficient to discharge output parasitic capacitances
107
and
108
of switches
103
and
104
to achieve the ZVS condition, even at a small load current. However, parasitic capacitances
112
and
113
of leading-leg switches
101
and
102
are discharged by energy stored in leakage inductance
109
of transformer
105
. (During the switching of switches
101
and
102
, primary winding
105
a
is shorted by rectifiers
110
and
111
carrying the output current of filter inductor
106
.) Since leakage inductance
109
is small, switches
101
and
102
cannot achieve ZVS condition even at relatively high output currents. In the prior art, the ZVS range of leading-leg switches
101
and
102
is extended either by increasing leakage inductance
109
, or by adding an external inductor in series with primary winding
105
a
. A properly sized external inductor can store enough energy to achieve ZVS condition in the leading-leg switches
101
and
102
even at low currents. However, a large external inductor would also store a large amount of energy at the full load, thus producing a large circulating energy adversely stressing the semiconductor components and reducing conversion efficiency. Further, in converter
100
, severe parasitic ringing may occur in the secondary winding
105
b
of transformer
105
when one of rectifiers
110
and
111
turns off. Such ringing results from a resonance among the junction capacitance of the rectifier, leakage inductance
109
and the external inductor (when present). To control such ringing, a snubber circuit is required on the secondary side of transformer
105
, thus significantly lowering the conversion efficiency of the circuit.
Alternatively, in the prior art, the ZVS range of switches
101
and
102
is extended to lower load currents without a significant increase of the circulating energy by using a saturable external inductor, as illustrated by full-bridge ZVS PWM converter
200
of FIG.
2
. (In this discussion and in the detailed description below, to facilitate correspondence between figures, like elements are assigned like reference numerals). Converter
200
is described in the article, “An Improved Full-Bridge Zero-Voltage-Switched PWM Converter Using a Saturable Inductor,” by G. Hua, F. C. Lee, M. M. Jovanovic, published
IEEE Power Electronics Specialists' Conf: Rec.,
pp. 189-194, 1991. When saturable inductor
209
is sufficiently large to saturate at a high load current, a controlled amount of energy is stored in saturable inductor
209
. At the same time, at a low load current (i.e., when saturable inductor
209
is not saturated), saturable inductor
209
has a sufficiently high inductance to store enough energy to provide ZVS in switches
101
and
102
even at small loads. However, when placed in the primary side of transformer
201
, saturable inductor
209
requires a relatively large magnetic core, thus increasing the cost of converter
200
. (Generally, a large magnetic core is required to eliminate excessive heat resulting from core loss as the flux in a saturable inductor swings between the positive and negative saturation levels).
In the prior art, the ZVS range of a FB ZVS PWM converter is also extended to a lower load current by placing saturable inductors on the secondary side, as illustrated by FB ZVS PWM converter
300
of FIG.
3
. As shown in
FIG. 3
, saturable inductors
309
a
and
309
b
are connected in series with rectifiers
110
and
111
, so that the flux swing in each of saturable inductors
309
a
and
309
b
is confined between zero and a positive saturation level (i.e., the flux swing in each of saturable inductors
309
a
and
309
b
is approximately half the flux swing of saturable core
209
of
FIG. 2.
) As a result, core loss in converter
300
in
FIG. 3
is reduced, as

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